Charge packet signal processing using pinned photodiode devices

ABSTRACT

An image sensor may include an array of image pixels coupled to analog-to-digital conversion circuitry formed from pinned photodiode charge transfer circuits. Majority charge carriers for the pinned photodiodes in the charge transfer circuits may be electrons for photodiode wells formed from n-type doped regions and may be holes for photodiode formed from p-type doped regions. Pinned photodiodes may be used for charge integration onto a capacitive circuit node. Pinned photodiodes may also be used for charge subtraction from a capacitive circuit node. Comparator circuitry may be used to determine digital values for the pixel output levels in accordance with single-slope conversion, successive-approximation-register conversion, cyclic conversion, and first or second order delta-sigma conversion techniques. The array of image pixels used for imaging may have a conversion mode wherein at least a portion of the pixel circuitry in the array are operated similar to the charge transfer circuits.

BACKGROUND

This relates generally to imaging systems and, more particularly, tosignal processing circuitry that utilizes pinned photodiode devices fordelivering charge to a circuit node.

Modern electronic devices such as cellular telephones, cameras, andcomputers often include camera modules having digital image sensors. Animage sensor (sometimes referred to as an imager) is formed from atwo-dimensional array of image sensing pixels. Each pixel receivesincident photons (light) and converts the photons into electricalsignals.

Capturing images using an image sensor involves using reading out pixelsignals from a subset of pixels from the two-dimensional image sensingpixel arrays (sometimes referred to as a “readout operation” of an imagesensor). Pixel signals may be routed or otherwise provided to signalprocessing circuitry during the readout operation. A readout operationmay be said to conclude when the signal processing circuitry thatreceives the image pixel signals converts the image pixel signals todigital image data. Prior to the read out of pixel signals from a subsetof the pixels in an array, the reset levels from the subset of thepixels in the array are also read out and converted to digital resetlevel data by the signal processing circuitry on the image sensor.

Converting pixel reset levels and pixel signals from analog signals todigital data is accomplished by analog-to-digital converter (ADC)circuitry. Conventional ADC circuits sometimes utilizepoly-insulator-poly or metal-insulator-metal capacitors having largesubstrate area requirements, density requirements, linearityrequirements, and extra silicon processing steps to form them.Capacitors may be used in switch capacitor circuits that providereference charges to the comparator circuitry in the ADC circuitry.Comparator circuitry in the traditional ADC circuitry itself oftenrequires capacitors. The capacitors in ADC circuitry are often used totransfer large amount of charges between nodes, resulting in excessivepower consumption and dissipation in the signal processing circuitry.Moreover, capacitors that are formed in signal processing circuitry arenot customizable as far as specialized silicon processing needed to meetcapacitor device performance specifications, thereby limiting theapplications and configurability of an image sensor that relies oncapacitors to provide references charges for an ADC circuit.

An image sensor that lacks silicon process customization orconfigurability for specialized analog circuit components in its imageprocessing circuitry cannot be optimized for particular applications asreadily, if at all, when compared to sensors having configurableprocessing circuitry. Furthermore, reliance on capacitors to transferlarge charge packets between nodes often results in excess powerconsumption and dissipation in the signal processing circuitry, furtherlimiting the applicability of the capacitor-based signal processingcircuitry to systems with larger and more costly power sources and heatdissipation capabilities that are suited to the power requirements ofthe capacitor-based circuitry. Capacitors used in ADC circuitry are alsoused to charge mixing, which occurs when capacitors are connectedtogether or coupled to a common node and settle to a common voltage. Acapacitor in signal processing circuitry with a charge level transferscharges to a second capacitor at a lower charge level when an electricalpath is formed between the two capacitors, resulting in a mixing ofcapacitor signals when the two capacitors are not electrically isolatedfrom one another like with an amplifier in a switched capacitor circuittopology.

It would therefore be desirable to provide improved signal processingcircuitry without reliance on conventional high performance capacitorsthat dissipate power to support charge mixing or dissipate power tosupport switched capacitor circuit topologies.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of an illustrative imaging system with animage sensor having image sensor pixels in accordance with anembodiment.

FIG. 2 is a simplified block diagram of an imager in accordance with anembodiment of the present invention.

FIG. 3A is a schematic of a single slope analog-to-digital converter(ADC) with pinned photodiode charge transfer circuits in accordance withan embodiment of the present invention.

FIG. 3B is a timing diagram for operating the single slope ADC withpinned photodiode charge transfer circuits of FIG. 3A in accordance withan embodiment of the present invention.

FIGS. 4A-4D illustrate steps for filling a pinned photodiode chargetransfer circuit and transferring charge packets to a summing node inaccordance with an embodiment of the present invention

FIG. 5 is a graph showing the generation of a ramp voltage using the ADCof FIG. 3 in accordance with an embodiment of the present invention.

FIG. 6A is a voltage level shifter that changes the polarity and voltagelevel of an input voltage in accordance with an embodiment of thepresent invention.

FIG. 6B is a graph showing the generation of a ramp voltage using asingle slope ADC without a comparator signal dependent offset inaccordance with an embodiment of the present invention.

FIG. 7A is a schematic of a single slope ADC without a comparator signaldependent offset, with pinned photodiode charge transfer circuits inaccordance with an embodiment of the present invention.

FIG. 7B is a timing diagram for operating the single slope ADC without acomparator signal dependent offset, with pinned photodiode chargetransfer circuits of FIG. 7A in accordance with an embodiment of thepresent invention.

FIG. 8A is a schematic of a charge cell basedsuccessive-approximation-register (SAR) ADC with pinned photodiodecharge transfer circuits in accordance with an embodiment of the presentinvention.

FIG. 8B is a schematic of a voltage level shifter that changes thevoltage level of an input voltage in accordance with an embodiment ofthe present invention.

FIG. 8C is a graph showing the voltages generated at summing nodes inthe SAR ADC of FIG. 8A in accordance with an embodiment of the presentinvention.

FIG. 8D is a timing diagram for operating the charge cell based SAR ADCwith pinned photodiode charge transfer circuits of FIG. 8A in accordancewith an embodiment of the present invention.

FIG. 8E is a flowchart of steps for setting the bits and operating thepinned photodiode based charge transfer circuits of the SAR ADC of FIG.8A in accordance with an embodiment of the present invention.

FIG. 9 is a schematic of a charge cell based SAR ADC with pinnedphotodiode charge transfer circuits and an auto-zero comparator inaccordance with an embodiment of the present invention.

FIG. 10 is a schematic of a charge cell based SAR ADC with pinnedphotodiode charge transfer circuits and an auto-zero comparator thatreceives a pixel voltage level directly from a pixel in accordance withan embodiment of the present invention.

FIG. 11 illustrates a charge cell based comparator that can be used withthe SAR ADC of FIG. 10 in accordance with an embodiment of the presentinvention.

FIG. 12 illustrates a charge cell based comparator that includesfloating gate transistors that can be used with the SAR ADC of FIG. 10in accordance with an embodiment of the present invention.

FIG. 13A is a schematic of a first order delta-sigma ADC withelectron-based and hole-based pinned photodiode structures in accordancewith an embodiment of the present invention.

FIG. 13B is a block diagram showing the functional blocks of the firstorder delta-sigma ADC of FIG. 13A in accordance with an embodiment ofthe present invention.

FIG. 13C is a timing diagram for operating the first order delta-sigmaADC with electron-based and hole-based pinned photodiode structures ofFIG. 13A in accordance with an embodiment of the present invention.

FIG. 14 is a schematic of a first order delta-sigma ADC withouthole-based pinned photodiodes in accordance with an embodiment of thepresent invention.

FIG. 15 is a schematic of a first order delta-sigma ADC of FIG. 14 thathas a constant pinning voltage level provided to all of the pinnedphotodiode devices in accordance with an embodiment of the presentinvention.

FIG. 16A is a schematic of a second order delta-sigma ADC withelectron-based and hole-based pinned photodiode structures in accordancewith an embodiment of the present invention.

FIG. 16B is a block diagram showing the functional blocks of the secondorder delta-sigma ADC of FIG. 16A in accordance with an embodiment ofthe present invention.

FIG. 16C is a timing diagram for operating the second order delta-sigmaADC with electron-based and hole-based pinned photodiode structures ofFIG. 16A in accordance with an embodiment of the present invention.

FIG. 16D is a schematic of a second order delta-sigma ADC withelectron-based and hole-based pinned photodiode structures with improvedsignal range in accordance with an embodiment of the present invention.

FIG. 16E is a timing diagram for operating the second order delta-sigmaADC with electron-based and hole-based pinned photodiode structures ofFIG. 16D in accordance with an embodiment of the present invention.

FIG. 17A is a schematic of a cyclic ADC with pinned photodiode chargetransfer circuits in accordance with an embodiment of the presentinvention.

FIG. 17B is a timing diagram for operating the cyclic ADC of FIG. 17A inaccordance with an embodiment of the present invention.

FIG. 18 is a schematic and timing diagram for a pre-emphasis circuit forcompensating for non-linear signal outputs over an input signal range inaccordance with an embodiment of the present invention.

FIG. 19 is a schematic of a pre-emphasis circuit for adding positive ornegative compensation signals to an ADC capacitive node in accordancewith an embodiment of the present invention.

FIG. 20 is a schematic of pixel groups in an imaging array that can beselectively used in a conversion mode in accordance with an embodimentof the present invention.

DETAILED DESCRIPTION

Embodiments of the present invention relate to signal processingcircuitry configured to transfer charge packets having an adjustablesize to a circuit node. Adjustable size charge packets may originate atpinned photodiode structures. Adjustable size charge packets may betransferred to circuit nodes that provide a reference voltage for acomparator in a signal processing circuit such as an ADC.

An electronic device with a digital camera module is shown in FIG. 1.Electronic device 10 may be a digital camera, a computer, a cellulartelephone, a medical device, or other electronic device. Camera module12 (sometimes referred to as an imaging device) may include image sensor14 and one or more lenses 28. During operation, lenses 28 (sometimesreferred to as optics 28) focus light onto image sensor 14. Image sensor14 includes photosensitive elements (e.g., pixels) in whichphotogenerated charges are produced in response to the light incident tothe pixels. Image sensors may have any number of pixels (e.g., hundreds,thousands, millions, or more). A typical image sensor may, for example,have millions of pixels (e.g., megapixels). As examples, image sensor 14may include bias circuitry (e.g., source follower load circuits), sampleand hold circuitry, correlated double sampling (CDS) circuitry,amplifier circuitry, analog-to-digital (ADC) converter circuitry, dataoutput circuitry, memory (e.g., buffer circuitry), address circuitry,etc.

Still and video image data from image sensor 14 may be provided to imageprocessing and data formatting circuitry 16 via path 26. Imageprocessing and data formatting circuitry 16 may be used to perform imageprocessing functions such as automatic focusing functions, depthsensing, data formatting, adjusting white balance and exposure,implementing video image stabilization, face detection, etc.

Image processing and data formatting circuitry 16 may also be used tocompress raw camera image files if desired (e.g., to Joint PhotographicExperts Group or JPEG format). In a typical arrangement, which issometimes referred to as a system on chip (SOC) arrangement, camerasensor 14 and image processing and data formatting circuitry 16 areimplemented on a common integrated circuit. The use of a singleintegrated circuit to implement camera sensor 14 and image processingand data formatting circuitry 16 can help to reduce costs. This is,however, merely illustrative. If desired, camera sensor 14 and imageprocessing and data formatting circuitry 16 may be implemented usingseparate integrated circuits.

Camera module 12 may convey acquired image data to host subsystems 20over path 18 (e.g., image processing and data formatting circuitry 16may convey image data to subsystems 20). Electronic device 10 typicallyprovides a user with numerous high-level functions. In a computer oradvanced cellular telephone, for example, a user may be provided withthe ability to run user applications. To implement these functions, hostsubsystem 20 of electronic device 10 may include storage and processingcircuitry 24 and input-output devices 22 such as keypads, input-outputports, joysticks, and displays. Storage and processing circuitry 24 mayinclude volatile and nonvolatile memory (e.g., random-access memory,flash memory, hard drives, solid state drives, etc.). Storage andprocessing circuitry 24 may also include microprocessors,microcontrollers, digital signal processors, application specificintegrated circuits, or other processing circuits.

FIG. 2 illustrates a simplified block diagram of imager 200 (e.g., animage sensor such as image sensor 14 of FIG. 1). Pixel array 201includes a plurality of pixels containing respective photosensitiveelements or regions arranged in a predetermined number of columns androws. The row lines that are coupled to the pixels may be selectivelyactivated by row driver 202 in response to row address decoder 203 andthe column select lines may be selectively activated by column driver204 in response to column address decoder 205. Thus, a row and columnaddress may be provided for each pixel. Row driver 202 and column driver204 may be activated in accordance with electronic rolling shutterreadout methods, or global shutter readout methods in imagers 200 thatsupport rolling shutter readouts.

Imager 200 is operated by a timing and control circuit 206, whichcontrols decoders 203, 205 for selecting the appropriate row and columnlines for pixel readout, and row and column driver circuitry 202, 204,which apply driving voltages to the drive transistors of the selectedrow and column lines. The pixel signals, which typically include a pixelreset signal Vrst& and a pixel image signal Vsig for each pixel aresampled by sample and hold circuitry 207 associated with the columndriver 204. A differential signal Vrst-Vsig is produced for each pixel,which is amplified by amplifier 208 and digitized by analog-to-digitalconverter 209. The analog to digital converter 209 converts the analogpixel signals to digital signals, which are fed to image processor 210which forms a digital image.

Analog-to-digital converter 209 may, in contrast to conventionalcapacitor-based ADC circuits, utilize pinned photodiodes to supply fixedamounts of charge into circuits that perform analog to digitalconversion (ADC) as well as other circuit functions. Pinned photodiodesmay be used for complete charge transfer to another node. Conventionalcapacitor circuits fail to achieve these features of completelytransferring fixed amounts of charges from one node to another, at leastbecause when transferring charges from one capacitor to another, thecompleteness of the charge transfer during a charge transfer interval isdependent on the relative voltage or charge levels of the twocapacitors. When, for example, it is desirable to transfer the chargesfrom a first capacitor or capacitive node to a second capacitor orcapacitive node, and when the first capacitive node is at a lowervoltage level than the second capacitive node, then the desired transfermay be unachievable without using a switch capacitor amplifier topologyin traditional circuitry where an electrical connection between thefirst and second capacitive nodes is relied upon to transfer thecharges.

Moreover, when the charge from a first capacitor is to be transferred toa second capacitor that has an existing charge or voltage level, chargeswill flow between the two capacitors when they are electricallyconnected, resulting in a mixing of charges from the two capacitors ateach of the first and second capacitors. Notably, such a transferbetween capacitors, in which charges are mixed, results in charges thatwere originally present at the second capacitor to be lost, ortransferred to the first capacitor which was only intended to be asource of charges but not a destination, or sink, for the charges.

However, pinned photodiodes, may be used to achieve one-directionalcharge transfer to any circuit node. Pinned photodiodes may include aphotosensitive region such as a photodiode region that is provided witha pinning layer with a built-in bias at a pinning voltage level Vpinthat is determined by doping levels. Complete, one-directional chargetransfer from a pinned photodiode to a circuit node may be possible whenthe circuit node has a potential at a level that is higher than the Vpinlevel for electron based pinned photodiodes.

Photodiode regions for charge collection of photon generated charge inthe pinned photodiodes may be n-type regions or p-type regions formed ina semiconductor substrate below the surface (sometimes referred to as“buried” below the surface or “buried in the substrate). Surface pinningregions formed over the photodiode regions (sometimes referred to as“well regions” or “photodiode wells”) of pinned photodiodes may be dopedwith dopants of an opposite dopant type than the photosensitive wellregions themselves in order to generate a charge collection area with aspecific built-in Vpin potential to hold charge. As an example, a p-typeor p+ surface pinning layer may be formed over an n-type photodiode wellregion. Similarly, an n-type or n+ surface pinning layer may be formedover a p-type photosensitive region. Pinned photodiodes can eithersubtract or add charge (or in another view, voltage inverselyproportional to node capacitance) from a node, based on the dopant typeassociated with the photodiode regions of the pinned photodiodes.

Specifically, when a pinned photodiode is formed with an n-typephotodiode region, the majority charge carriers in the n-type photodioderegion will be electrons. When charges in the pinned photodiode having an-type photodiode region are transferred to another circuit node,electrons are transferred to the another circuit node, thereby addingthe electron charges to the another circuit node. Transferring electronsfrom the pinned photodiode to a circuit node may reduce the voltage atthe circuit node by an amount proportional to the number of electronsthat were transferred to the circuit node.

Similarly, in pinned photodiodes having a p-type photodiode region, themajority charge carriers in the p-type photodiode region are “holes,”which may act as if they have properties associated with positivecharges. When the holes from a p-type photodiode region of a pinnedphotodiode are transferred to a circuit node, the addition of holes tothe circuit node effectively remove electrons from the circuit node,which may thereby reduce the amount of charges on the circuit node, andwhich may increase the voltage at the circuit node.

Pinned photodiodes having n-type photodiode regions that transfercharges (specifically electrons, sometimes denoted as “e-”) aresometimes referred to (in this disclosure) as “charge integrators,” andpinned photodiodes having p-type photodiode regions that transfer“holes” and thereby remove electrons are sometimes are sometimesreferred to as “charge subtractors.” Pinned photodiodes havingphotodiode regions of either type may transfer charges in packetscorresponding to a full well capacity of the photodiode regions in thepinned photodiodes. Compared to conventional signal processing circuitryusing capacitor circuits to transfer charges, the relatively smallersize of charge packets that are transferred from pinned photodiodes mayallow for lower power circuits, relative to traditional capacitor-basedcircuits.

FIG. 3 illustrates a single-slope analog-to-digital converter (ADC)circuit in accordance with an embodiment. Analog-to-digital converter300 may receive analog pixel signals from an image pixel in an imagesensor pixel array. ADC converter 300 may include multiple transistorsand pinned photodiode circuits that include one or more respective dopedregions in a semiconductor substrate. The dopants used in the varioustransistor and pinned photodiode devices may be reversed, relative tothe exemplary embodiments described below. As an example, when circuitryis described as including a transistor that, in the exemplaryembodiment, is a p-channel or p-type transistor, and a pinned photodiodethat, in the exemplary embodiment, has an n-type photodiode region forelectron collection with a surface p-type pinning layer, it should beappreciated that the circuitry can alternatively be implemented with thetransistor being an n-channel or n-type transistor and with the pinnedphotodiode having a p-type photodiode region for hole collection with asurface n-type pinning layer. When the dopant regions of the devices inthe circuitry are “reversed” in this way, supply voltages and controlsignals may also be adjusted to suit the properties of the devices withthe “reversed” dopant regions.

A pixel supply voltage VAA may be provided to the ADC converter 300 at asupply terminal 302. VAA may be a positive pixel supply voltage. Apre-charge transistor 304 may selectively couple the supply voltageterminal 302 to a floating node 308. Pre-charge transistor 304 may be ap-channel transistor (such as a planar PMOS transistor) having a gatethat is asserted when a logic “low” or ground voltage is applied to thegate of the transistor 304. Node 308 may be referred to as floatingbecause it may not be connected, at least not constantly, to any voltagesource. Therefore, the floating node 308 may be effectively isolatedfrom other nodes in the converter 300, in that charges on the floatingnode 308 may remain stable when the control signals to the varioustransistors, that result in the connection of the floating node 308 toany other node, are de-asserted.

Pinned photodiode devices PPD_1 in ADC converter 300 may correspond ton-type photodiode regions provided with a p-type pinning layer asdescribed above. In other words, the PPD_1 pinned photodiodes may beelectron accumulation devices having n-type photodiode well regions,with a p-type pinning layer that is formed over the photodiode wellregion and inherently contributes to setting a constant built-in pinningvoltage determined by doping levels in the n-type photodiode, surfacep-type pinning layer, and surrounding p-type substrate. The built-inpinning voltage Vpin (not marked in FIG. 3) of the PPD_1 pinnedphotodiodes may determine the conditions in which a full charge transferfrom any one of the PPD_1 pinned photodiodes is possible. Specifically,when a node such as floating node 308 has a voltage potential level thatis greater than the pinning voltage Vpin of a given pinned photodiodePPD_1, a complete transfer of the electrons from the PPD_1 node to thefloating node 308 may be possible. In the descriptions below, it will beassumed that the pinning voltages at pinned photodiode devices of eitherdopant type (i.e., pinned photodiodes with n-type photodiode regions andp-type pinning layers or pinned photodiodes with p-type photodioderegions and n-type pinning layers) are constructed with a suitablebuilt-in voltage at their pinning layer that enables the completetransfer of charges from the pinned photodiodes to the node at which thecharges are to be received.

As illustrated in FIG. 3, multiple pinned photodiodes PPD_1 may beprovided in parallel charge transfer circuits 340. In the most basicoperating mode of the ADC circuitry 300, only a single charge transfercircuit 340 is required. However, the inclusion of multiple parallelcharge transfer circuits 340 (sometimes referred to herein as “chargetransfer stages” 340) may improve the speed at which the ADC circuit 300can operate, by enabling a faster integration of charges on a floatingnode 308 through the multiple charge transfers from each of the parallelcharge transfer stages 340 to the floating node 308. Charge transfers toand from the multiple parallel charge transfer stages 340 may besimultaneous. Or, each of the parallel charge transfer stages 340 may beindependently controllable to transfer charges to and from itsrespective pinned photodiode PPD_1, regardless of the control signalsapplied to any other charge transfer stage 340.

In certain embodiments, it may be desirable to group sets of parallelcharge transfer stages 340 and provide the groups of parallel chargetransfer stages 340 with the same control signals. As an example, if anADC circuit 300 includes 64 parallel charge transfer stages 340, eightgroups of 8 parallel charge transfer stages 340 may be formed, with eachgroup of parallel charge transfer stages 340 receiving common controlsignals which effectively operate each of the charge transfer stages 340simultaneously, and in an identical manner. Parallel charge transferstages 340 may, in this way, be divided into one, two, or any number ofgroups.

Each charge transfer stage 340 may include a fill transistor 332 thatconnects a ground supply terminal 306 to the pinned photodiode PPD_1when the fill transistor 332 gate is asserted, thereby turning on thefill transistor 332. Ground supply terminal 306 may provide a constantsupply voltage of 0V, or any other suitable voltage. Ground supplyterminal 306 may be an adjustable supply terminal whose voltage can beadjusted during operation of the ADC 300. Fill transistors 332 may beviewed as transfer gates between the ground supply terminal 306 and thepinned photodiodes PPD_1 in their respective charge transfer stages 340.Fill transistors 332 may be formed in a manner similar to ananti-blooming gate is formed relative to the pinned photodiodes in theimage pixel array (not shown).

Activation of the fill transistors 332 may cause charges to accumulatein the pinned photodiodes PPD_1. In the exemplary embodiment of FIG. 3,where the pinned photodiodes PPD_1 have n-type photodiode wells formedwith p-type pinning layers over the n-type photodiode wells, activationof the fill transistors 332 may cause electrons to accumulate in thepinned photodiodes PPD_1. The amount of charge transferred into thePPD_1 when the fill transistor is activated may be based on the durationof the interval during which the fill transistor 332 is activated, thevoltage level on the fill transistor drain, and on the full wellcapacity of the pinned photodiode that is determined by the doping levelof the photodiode.

Each charge transfer stage 340 may also include a transfer transistor334 that connects the pinned photodiode PPD_1 to the floating node 308(sometimes referred to herein as the Cdac1 node) when the transfertransistor 334 gate is asserted, thereby turning on the transfertransistor 334. Asserting the transfer transistor 334 may allow thecharges accumulated in the pinned photodiode PPD_1 to be completelytransferred to the floating node 308.

In a given charge transfer stage 340, the respective pinned photodiodePPD_1, the fill transistor 332, and transfer transistor 334 are ideallythe same devices as used in the image pixel array from which the ADC 300receives pixel reset and pixel signal levels. Fill transistor 332transistor may correspond to the anti-blooming AB gate for a pixel whichdrains excess charges in a pixel photodiode to a supply voltage toprevent excess charges from contaminating other nodes in the pixel orneighboring pixels. Transfer transistor 334 may correspond to the pixeltransfer gate which transfers accumulated photogenerated charges fromthe pinned photodiode of an image pixel to a pixel floating diffusionnode.

In this way, pixel structures that are already designed and optimizedfor use in the image sensor pixels that produce image signals may beleveraged elsewhere on the image sensor die (or on a separate die, in astacked-die embodiment) in processing circuitry such as the ADC 300.Implanting the various regions of semiconductor substrate to produce thetransistors and pinned photodiodes in the charge transfer stages 340 maybe performed using substantially the same methods by which the dopedregions for corresponding structures in the pixels of the image pixelarray are formed and implanted.

The floating node 308 may be used to generate a single slope ramp foruse in the operation of ADC 300. This ramp values generated at thefloating node 308 may be compared to the pixel value stored on theVref_comp capacitor 312. Specifically, a pixel sample-and-hold output316 from the image pixel array may be transferred via a pixel outputtransfer transistor 314. Transistor 314 may be used to transfer either apixel signal level corresponding to an amount of charge from thephotodiode of a given pixel in the image pixel array, or a pixel resetlevel corresponding to an output level of a pixel that has been reset.Pixel reset levels may be converted using ADC 300 for use in correlateddouble sampling (CDS) imaging. Both pixel reset and pixel signal levelsare digitized and subsequent digital CDS circuitry (not shown) may beused to generate a final pixel digital value, often by subtracting thepixel reset level from the pixel signal level.

Prior to the conversion operations of ADC 300, the pixel row select maybe activated at time t1 of FIG. 3B, which produces a correspondingsignal at the pixel output received by the sampling transistor 314. Att2 of FIG. 3B subsequent to t1, the pixel reset signal is asserted,causing the pixel output to reflect the pixel reset level.

The ADC 300 operation may begin with the Cdac1 node 308 beingpre-charged from the supply terminal 302 via the pre-charge transistor304, to a pre-charge voltage. The pre-charge operation may correspond tothe asserted signal on the pre-charge line between t3 and t4 of FIG. 3B,during which the PMOS transistor 304 gate signal is deasserted toprecharge the Vdac1 node (i.e., event 1802 of FIG. 3B). The pre-chargevoltage may be any level, but for the purposes of illustration will beassumed to be 2.8V. Floating node 308 is pre-charged while thecomparator 318 is auto-zeroed (also at time t3, until time t4 of FIG.3B), when applicable based on the specific implementation of ADC 300,while the input capacitors 316-1 and 316-2 are clamped (i.e., theclamping switches 314-1 and 314-2 are closed to connect the Vclampsupply voltage terminals to the capacitors 316), and while the pixelreset level is sampled on the Vref_comp capacitor 312 (i.e., when areset level provided at the pixel output 316 is transferred to theVref_comp capacitor 312 via the pixel output transfer transistor 314).The sampling of the pixel reset level may occur in the interval betweent3 and t5 when the SHR signal provided to the sampling transistor 314 isasserted as shown in FIG. 3B. After the pre-charge operation, the PPD_1device in at least one charge transfer stage 340 may be filled withelectrons, in the most basic mode of operation. FIGS. 4A-4D arepotential diagrams (sometimes referred to as a “fluid diagrams”) thatillustrate how a pinned photodiode PPD_1 may be filled with electronswhich are then transferred to a floating circuit node, in accordancewith an embodiment. Turning to FIG. 4A, a generic charge transfer stage340 is illustrated with the fill transistor 332 being represented by thefill gate 432. The pinned photodiode PPD_1 may be filled up withelectrons by keeping the drain of the fill transistor 332 at a groundvoltage (such as 0V, for example) while asserting the gate of the filltransistor 332 to turn the transistor on, as is illustrated at time t6of FIG. 3B where the Fill1 signal provided to the fill transistor 332-1in the charge transfer stage 340-1 is asserted. In FIG. 4A, the fillgate 432 is asserted or activated, allowing the supply voltage electrons492 in the supply region 406 of the diagram (corresponding to the groundsupply 306 of FIG. 3), to flow through the fill transistor (i.e.,through the channel below the fill gate 432) and into the pinnedphotodiode PPD_1 well region 436. Because the transfer transistor 334(shown in FIG. 4A as transfer transistor gate 434) is de-asserted,electrons in the pinned photodiode PPD_1 are unable to travel throughthe channel below the transfer transistor gate 434 into the floatingnode 310 (shown in FIG. 4A as Cdac1 well region 410).

FIG. 4B illustrates the subsequent state of the charge transfer stage340, specifically illustrating the fill gate 432 being de-asserted (suchas at t7 of FIG. 3B where the Fill2 signal provided to the filltransistor 332-1 in the charge transfer stage 340-1 is de-asserted), inwhich the pinned photodiode PPD_1 well may be isolated from the supplyregion 406 (whose shaded charges are not illustrated in FIGS. 4B-4D, soas to avoid unnecessarily obscuring the relevant features of thedrawings). In FIG. 4B, the pinned photodiode PPD_1 may be filled to itsfull-well capacity, or the maximum amount of majority charge carriersthat can be stored in the n-type photodiode well of the pinnedphotodiode PPD_1. In the exemplary embodiments described herein, it maybe assumed that the full well capacity of pinned photodiode PPD_1 is5,000 electrons. However, variations in the full well capacity of apinned photodiode PPD_1 may be determined by the formation of the pinnedphotodiode structure during silicon processing. Moreover, the thermalenergy of electrons flowing through the channel under the filltransistor gate 432 of the fill transistor 332 introduces thermal noise(sometimes referred to herein as “kTC noise”) which may cause the exactamount of charge or the exact number of electrons in the “filled” pinnedphotodiode PPD_1 of FIG. 4B to vary. FIG. 4C illustrates how, when thegate 434 of a transfer transistor 334 in a charge transfer stage 340 isasserted, turning the transfer transistor 334 on, the charges 494 thatfilled up the pinned photodiode PPD_1 are transferred to the floatingnode 308 well region 410 (such as at time t8 of FIG. 3B, where the TX1signal provided to the transfer transistor 334-1 in charge transferstage 340-1 is asserted). Charges 496-1 may flow from the pinnedphotodiode PPD_1 well region 436 across the channel under the transfertransistor 334 gate 434 into the floating node 308 well region 410.Charges 496-2 that are transferred into the floating node 308 wellregion 410 may be prevented from transferring back into the PPD_1 wellby the potential barrier between the floating node 308 well region 410and the pinned photodiode well region 432, even when the transfertransistor gate 434 is asserted as illustrated in FIG. 4C.

FIG. 4D illustrates how, subsequent to the complete charge transfer ofFIG. 4C, the transfer transistor 334 gate 434 may be deassserted at timet9 of FIG. 3B where the TX1 signal provided to the transfer transistor334-1 in charge transfer stage 340-1 is de-asserted, with thetransferred charges 498 corresponding to a full-well capacity of thepinned photodiode PPD_1 are transferred or inserted into the floatingnode 308 well region 410. Because the full-well capacity of the pinnedphotodiode corresponds is used in each full-transfer of a chargetransfer stage 340, each full-well capacity of a pinned photodiode PPD_1may be treated as a charge packet. Transferring a charge packet from apinned photodiode PPD_1 of a single charge transfer stage 340 to thefloating node 308 may cause the voltage at the floating node 308 to dropor decrease by 0.8 mV from its previous voltage, assuming a full wellcapacity of 5,000 electrons and a 1 picofarad capacitance of floatingnode 308 (calculated by solving for V using the aforementioned C=Q/Vequation). In ADC circuitry that uses electron accumulation pinnedphotodiodes (i.e., photodiodes that have an n-type photodiode well),counter 320 may decrement a count value, because the voltage at thefloating Cdac1 node 308 with which the sampled voltage across Vref compcapacitor 312 is compared is a decreasing voltage. The voltage at thefloating Cdac1 node 308 decreases when the pinned photodiodes PPD_1 haven-type photodiode wells, because negative charge packets (i.e., packetsof electrons having a size corresponding to a full well capacity of oneor more photodiode wells) are successively transferred to the floatingCdac1 node 308 using the charge transfer stages 340.

The successive transfer of negative charge packets to the floating Cdac1node 308 may be used to generate a decreasing ramp signal that beginswith an initial value based on the initial pre-charge of the floatingCdac1 node 308 by the supply voltage 302 via the pre-charge transistor304. At each clock cycle, one or more charge transfer stages 340 may beused to transfer negative charge packets to the floating Cdac1 node 308.The counter 320 may, at each cycle of the clock provided at the clockinginput of counter 320, decrement the counter from a maximum valuecorresponding to the bit-resolution of the ADC 300. At each clock cycle,a charge transfer stage 340 may be used to transfer a negative chargepacket corresponding to a full photodiode PPD_1 well to the floatingCdac1 node 308. In this way, at each clock cycle the voltage of thefloating Cdac1 node 308 may be lowered by an amount based on a pinnedphotodiode PPD_1 full well capacity, as the count maintained by thecounter 320 is decremented by one.

As mentioned previously, when counter 320 is implemented as adecrementing circuit, the initial value from which the counter 320begins decreasing corresponds to the bit-resolution of the ADC 300. Asan example, when the counter 320 is implemented as a decrementingcircuit in a 10-bit ADC, the initial value from which the counter 320starts decrementing is 1024 (corresponding to the maximum range ofdecimal values that can be represented by 10-unsigned data bits).Similarly, when the counter 320 is implemented as a decrementing circuitin a 12-bit ADC, the initial value from which the counter 320 startsdecrementing is 4096 (corresponding to the maximum range of decimalvalues that can be represented by 12-unsigned data bits). The countervalue maintained by the counter 320 may be provided at the output 326 ofthe counter 320, as an n-bit value in an ADC 300 with an n-bitresolution.

In certain embodiments, such as when the pinned photodiodes PPD_1 areimplemented as p-type wells with an n-type pinning layer, the chargesaccumulated in the pinned photodiodes PPD_1 with the p-type wells may beholes, which, when transferred to the floating Cdac1 node 308, mayincrease the voltage at the floating Cdac1 node 308. Floating Cdac1 node308 may have an effective capacitance illustrated by capacitor 310. Whenpinned photodiodes PPD_1 are implemented as p-type wells with an n-typepinning layer, the counter 320 may be configured to function as anincrementing circuit, because the successive transfers of positive(hole) packets to the floating Cdac1 node 308 may generate an increasingramp voltage at the floating Cdac1 node 308. Counter 320 may, at everyclock cycle, increment a count value starting from zero, as long as itis enabled. When pinned photodiodes PPD_1 are implemented with p-typewells, the enabling input of counter 320 may be produced by a comparatorthat is enabled when the voltage at the floating Cdac1 node 308 (atwhich an voltage ramp that increases every clock cycle is produced, bythe transfer of hole charge packets from the p-type well pinnedphotodiodes PPD_1) is less than the voltage across the Vref_compcapacitor 312. In other words, when the pinned photodiodes PPD_1 areformed with p-type wells that are filled with holes, which aretransferred to the floating Cdac1 node 308 using a transfer transistor334 in a charge transfer stage 340, the comparator 318 may have anegative input coupled to the coupling capacitor 316-1 illustrated inFIG. 3, and a positive input coupled to the coupling capacitor 316-2illustrated in FIG. 3.

Returning to the exemplary embodiment illustrated in FIG. 3, where thepinned photodiodes PPD_1 in the charge transfer stages 340 are used totransfer electron packets to the floating Cdac1 node 308, a voltage rampthat decreases from a pre-charge voltage every clock cycle by an amountcorresponding to the number of charge packets that are used to transfercharges to the floating Cdac1 node 308 may be generated at the floatingCdac1 node 308. FIG. 5 illustrates an exemplary decreasing voltage ramp510 that may be generated at the floating Cdac1 node 308. V1 in FIG. 5may correspond to the voltage level at floating Cdac1 node 308 after thegate of pre-charge transistor 304 is asserted, turning on pre-chargetransistor 304 and thereby charging the floating Cdac1 node 308 to thesupply voltage VAA 302 at time t1. For the purposes of illustration, VAAis illustrated in FIG. 5 as being 2.8 V, however any suitable voltagefor VAA may be used.

At time t2, one or more charge transfer stages 340, each of which haverespective pinned photodiodes PPD_1 filled with electrons (correspondingto the state illustrated in FIG. 4B), may transfer their respective oneor more electron packets (sometimes referred to as negative chargepackets) to the floating Cdac1 node 308. At time t3, the negative chargepackets have been transferred to the floating Cdac1 node 308, and havereduced the voltage at the floating Cdac1 node 308 to a voltage V2 thatis less than the pre-charge voltage V1. The difference between V1 and V2may be determined by the number of charge transfer stages 340 that areused to transfer negative charge packets to the floating Cdac1 node 308at time t2. As an example, if only a single charge transfer stage 340 isused to transfer a single negative charge packet, the difference betweenV1 and V2 may correspond to the decrease in voltage caused bytransferring a number of electrons corresponding to a full-well capacityof the pinned photodiode PPD_1 of the single charge transfer stage tothe floating capacitive node 308. Specifically, because C=Q/V can bere-written as V=Q/C, where Q is the amount (or change in the amount) ofcharge in coulombs, V is the voltage level (or change in the voltagelevel), and C is the capacitance, the change in voltage at the floatingcapacitive node 308 that results from transferring a charge packet maybe described as (n×F_w×−Q_e)/C, where n is the number of charge transferstages 340 that are used to transfer negative charge packets to thefloating capacitive node 308, F_w is the number of electrons that can bestored in a pinned photodiode PPD_1 (sometimes referred to as thefull-well capacity of the pinned photodiode PPD_1), where Q_e is themagnitude of a single electron charge (approximately −1.6×10⁻¹⁹Coulombs), and where C is the capacitance of a node 308.

Using the same exemplary values as mentioned in the above examples,where F_w is 5,000, and C is 1 picofarad, when a single charge transferstage 340 is used to transfer charge to the floating node 308 (i.e.,when n is 1), the difference between V2 and V1 is approximately 0.8 mV(or, 8×10⁻⁴V). At time t4, one or more charge transfer stages 340 may beused to transfer additional negative charge packets to the floating node308. The number of charge transfer stages 340 used to transferadditional negative charge packets to the floating node 308 at time t4may be the same as, or may be different from, the number of chargetransfer stages 340 used to transfer negative charge packets to thefloating node 308 at time t2. In a preferred embodiment or bit-encodingscheme, where successive digital values (e.g., any first digital valueand a second digital value that is greater than the first digital valueby one) in the bit-encoding scheme correspond to analog voltage levelsseparated by a constant difference (or, step size), the number of chargetransfer stages 340 that are used to transfer negative charge packets attime t2 may be the same as the number of charge transfer stages 340 thatare used to transfer negative charge packets at time t4. At time t5, thetransfer of negative charge packets to the floating Cdac1 node 308 maybe complete and the voltage at the floating node 308 may have a levelV3, corresponding to the change in voltage produced at the floating node308 as a result of the negative charge packets that were transferred tonode 308 at time t4.

The interval between t1 and t3 may be equal to the interval between t3and t5 and may correspond to a clock period. The interval between t1 andt3, and the interval between t3 and t5 may be referred to as a chargetransfer interval. A clock having a period equal to charge transferinterval may be provided to the counter 320 at its clocking input 328,in a preferred embodiment.

Returning to the exemplary analog-to-digital conversion of a pixel resetlevel on the Vref_comp capacitor 312, the pixel reset level may berepresented by the sampled signal level 516 in FIG. 5. Generally, aslong as the negative ramp signal 510 generated on the floating Cdac1node 308 is above the sampled signal level 516, whether the sampledsignal level 516 is a pixel reset level or a pixel signal level, thefloating node 308 voltage level coupled through coupling capacitor 316-1that is provided at the positive input of comparator 318 may be greaterthan the sampled voltage across Vref_comp capacitor 312 coupled throughcoupling capacitor 316-2 that is provided at the negative input ofcomparator 318. Consequently, as long as the negative ramp signal 510generated on the floating Cdac1 node 308 is above the sampled signallevel 516, the comparator 318 output provided at the enable input ofcounter 320 may be at a logic high level (a positive voltage level, inthe present example), enabling the counter 320, which decrements a countvalue from an initial value (as discussed above, 1024 for a 10-bitresolution ADC 300, and 4096 for a 12-bit resolution ADC 300) at everycycle of the clock signal received at the clocking input 328 of counter320.

When the negative ramp signal 510 drops below the sampled signal level516, the voltage level at the positive input of the comparator 318 maydrop below the voltage level at the negative input of the comparator318, causing the output of comparator 318 that is provided to theenabling input of the counter 320 to flip from a logic high level to alogic low level (a ground voltage level, in the present example). Thetransition of the output of the comparator 318 from a logic high levelto a logic low level may disable the counter 320 and indicate that theconversion is complete. When counter 320 is disabled, the count valuemaintained in counter 320 may be maintained, and may correspond to adigital value that corresponds to the sampled signal level across theVref comp capacitor 312. When counter 320 is disabled, circuitrycontrolling the ADC 300 may detect that the conversion operation iscomplete and proceed to subsequent conversions.

With the sampling and conversion operation of the pixel reset signaldescribed above, this first conversion of the pixel reset signal levelcorresponds to converting the kTC noise associated with the Cdac1 node308, the kTC noise associated with the coupling capacitors 316, and thekTC noise associated with the Vref_comp sampling capacitor 312 as wellas the comparator 318 offset. For a subsequent, second conversion,floating Cdac1 node 308 may be pre-charged to the supply voltage 302level via the pre-charge transistor 304, but the clamp switches 314 arenot closed, as they were in the first conversion of the pixel resetsignal level prior to sampling the pixel reset level on the Vref_compcapacitor 312 and generating the ramp at floating Cdac1 node 308, toconnect respective ends of the coupling capacitors 316-1 and 316-2 tothe Vclamp voltage. Instead, the coupling capacitors 316-1 and 316-2 arenot clamped again after the conversion of the pixel reset signal, inorder to maintain the kTC noise of the coupling capacitors 316. Thesampling transistor 314 may then be used to sample a pixel signal levelonto the Vref_comp capacitor 312.

Because the coupling capacitor 314-2 that is connected to the Vref compcapacitor 312 is not clamped, the input to the negative input ofcomparator 318 may correspond to the difference between the reset andsignal levels. Hence, the second conversion of the pixel signalcorresponds to the conversion of the kTC noise associated with thefloating Cdac1 node 308, the kTC noise associated with the Vref_compcapacitor 312, the comparator offset, and difference between the pixelresent and pixel signal levels. Digital correlated double sampling (CDS)between the results of the first conversion of the pixel reset level andthe second conversion of the pixel signal level may then remove thecomparator offset and kTC noise associated with the coupling capacitors314.

The pixel signal level may be converted after time t10 of FIG. 3B in amanner similar to the conversion of the pixel reset level, at least inthat the sampled voltage 516 across the sampling capacitor 312 may becompared to the negative ramp voltage 510 that is generated at thefloating node 308, through the coupling capacitors 316-2 and 316-1, bycomparator 318, which enables the counter 320. At time t10 of FIG. 3B,the pixel transfer may be asserted, which causes the pixel output levelto decrease by an amount Delta V1 that is proportional to the signalcharge at the floating diffusion node of the pixel. When the SHS signalis asserted at time t11, the pixel signal output may be sampled at thesampling capacitor 312, which may cause the voltage level atComparator_VIN_N to drop by the amount Delta V1 starting at t11. Also att11, the PMOS pre-charge transistor 304 gate signal is deasserted (i.e.,precharge is deasserted at t11 of FIG. 3B) causing the voltage at Vdac1to be reset to the pixel supply level VAA. Interleaved charge transferoperations 1844 and 1846 may commence, causing the voltage at the Vdac1node 308, and consequently the Comparator_VIN_P node to drop, andcausing the output of the comparator to be at a logic high level,enabling counter 320.

While enabled, counter 320 may decrement a count value by one, from aninitial value corresponding to the bit-resolution of the ADC 300, atevery cycle of the clock provided at the clocking input 328 of thecounter 320. When the decreasing ramp voltage at the floating node 308is less than the sampled pixel signal voltage level across the Vref_comp312, the output of comparator 318 may transition from a logic high levelto a logic low level as shown by event 1806 of FIG. 3B, therebydisabling the counter 320 and signaling the completion of the pixelsignal voltage level conversion. The last value of the count valuemaintained by counter 320 before the counter 320 is disabled maycorrespond to a digital value corresponding to the analog voltage levelacross the Vref_comp capacitor 312. Using the exemplary diagram of FIG.5, the voltage difference between charge transfer intervals (i.e., theinterval between t1 and t3 and the interval between t3 and t5) maycorrespond to a common voltage difference between V1 and V2, and V2 andV3. This common voltage difference amount sets the LSB size for theramp. For a 10-bit ADC 300, the ramp covers an input range of 1023steps×0.8 mV, assuming that only one charge transfer stage 340 is usedto transfer a charge packet to the floating Cdac1 node 308 during acharge transfer interval, which is approximately 878 mV. Because of theuncertainty in charge written into the pinned photodiode devices PPD_1,there may be uncertainty in the maximum value of the ramp at the end ofany conversion. This uncertainty may be proportional to the square rootof the number of transfers and the random uncertainty in pinnedphotodiode charge. Specifically, this uncertainty may be equal to thekTC noise associated with the number of charge transfer stages 340 usedto transfer charge packets to the floating Cdac1 node 308 during acharge transfer interval, multiplied by the square root of the number oftransfers needed to generate a full ramp of voltage levels (whichcorresponds to the bit-resolution of the ADC 300).

As an example, the maximum ramp value uncertainty assuming a singlecharge transfer stage is used to transfer a charge packet to thefloating Cdac1 node 308 during a charge transfer interval for a 10-bitADC may be determined by multiplying the kTC noise of 1.44 μV associatedwith the 9 electron kTC noise of a single charge transfer by the squareroot of 1024 (because 1024 is the number of steps in a full voltage rampin a 10-bit ADC), or 46 μV. Similarly the maximum ramp value uncertaintyassuming 10 charge transfer stages are used to transfer charge packetsto the floating Cdac1 node 308 during a charge transfer interval for a10-bit ADC may be given by 4.6 μV (noise associated with the 28 electronkTC noise of 10 charge transfer stages' charge packet transfers)multiplied by the square root of 1024, or 146 μV. Similarly the maximumramp value uncertainty assuming 40 charge transfer stages are used totransfer charge packets to the floating Cdac1 node 308 during a chargetransfer interval for a 10-bit ADC may be given by 9.1 μV (noiseassociated with the 57 electron kTC noise of 40 charge transfer stages'charge packet transfers) multiplied by the square root of 1024, or 0.29mV.

Even if there are a few hundred electrons of noise in a charge transferstage's charge transfer operation and the final noise in the ramp valueis larger than the LSB of the ADC 300 at the end of the conversion, shotnoise in the pixel signal is still much larger (878mV/5000*sqrt(5000)=11.6 mV, assuming an imaging pixel full well of 5000electrons).

This analysis may also be applied to a 12-bit implementation of ADC 300.Because of additional charge transfers (4095 total charge transfers fora full ramp of voltages), the voltage swing increases on Cdac1. Becausethe minimum voltage on Cdac1 is 1.5V (set by the Vpin potential toguarantee complete charge transfer), the maximum DAC range is2.8V−1.5V=1.3V. To enable 4095 charge packet transfers, the Cdac1 valueis increased to 4 pF. The corresponding LSB ramp size is 0.2 mV (i.e., aquarter of the LSB ramp size used in embodiments with a 1 pF capacitor,assuming a full well capacity of 5,000 electrons) and the correspondingmax ramp voltage swing is 819 mV (i.e., 4,096×0.2 mV), when the Cdac1value is 4 pF and a 12-bit counter 320 is used in ADC 300.

The size of the Cdac1 capacitor can be reduced for 12-bit ADC operationif the amount of fill charge is reduced. The fill charge refers to theamount of charge filled in a pinned photodiode PPD_1 in any given chargetransfer circuit 340 during a charge transfer operation, and is based atleast in part by the voltage at the ground supply 306. Rather thanfilling the PPD_1 with electrons while the ground supply 306 ismaintained at 0V, ground supply 306 may be held at a higher voltage suchas 0.75V using the same charge transfer operations described inconnection with FIGS. 4A-4D. When the ground supply 306 is held at ahigher voltage such as 0.75V, the effective fill level of electrons inthe supply region 406 (i.e., the voltage level provided at the groundsupply 306 of FIG. 3) may be visualized as filling the supply region 406up to a voltage level 452 that results in the pinned photodiode PPD_1well region 436 being filled only up to the level 452, as opposed tobeing filled to capacity (i.e., up to the level 450) when the supplyvoltage 306 provides 0V.

By reducing the amount of charge that pinned photodiodes PPD_1 in thecharge transfer stages 340 are used to transfer in each transferoperation, the Cdac1 capacitor size may be reduced while maintaining theability to store the charge from the 4096 charge transfer operationsfrom the one or more charge transfer stages 340 during the operation ofa 12-bit embodiment of ADC 300. Ideally, the range of voltages that the12-bit ADC 300 can convert is the same as the range of voltages that the10-bit embodiment of ADC 300 can covert. Though the amount of chargetransferred by pinned photodiodes PPD_1 in each charge transferoperation of stages 340 may be reduced by increasing the voltageprovided at supply 306, the range of voltages that can be generated atthe floating node 308 (which at least partially determines the voltagesthat can be converted by the ADC 300) may be selected by the size chosenfor capacitor Cdac1 310. Because the size of Cdac1 capacitor 310determines the voltage produced by the charges transferred by stages 340at the floating node 308, the size of Cdac1 capacitor 310 may beincreased or decreased based on the desired range of voltages to beconverted by ADC 300, the voltage provided at supply 306 to the chargetransfer circuits 340, and the full-well capacity of the pinnedphotodiodes PPD_1 in the charge transfer circuits 340.

Variation in full well capacity of pinned photodiodes PPD_1 as a resultof process variation is fixed, and calibration may be used to compensatefor non-random variation in ADC LSB step size and Vref node acrosscapacitor 312 when matching multiple ADC circuits is needed. Inconnection with the illustrative ramp generation of FIG. 5, it wasmentioned that while only a single PPD device in a charge transfer stage340 may be used to generate the negative slope ramp, multiple pinnedphotodiodes (PPDs) in multiple charge transfer stages 340 may be used togenerate the negative slope ramp. In FIG. 5, during the intervalsbetween times t1 and t2, and between times t3 and t4, one or more of thecharge transfer stages 340 takes time to fill its respective PPD _1device and then transfers the charges to the Cdac1 node 308 in theintervals between times t2 and t3, and between times t4 and t5. Thedifference between voltages V1, V2, and V3 may be determined at least inpart by the number of charge transfer stages 340 that are used totransfer charges from their respective PPD _1 devices onto the floatingCdac1 node 308 at times t2 and t4.

However, multiple PPD _1 devices can be used in order to pipeline thefill operations and have a faster ramp, with shorter intervals betweenchanges in the generated ramp voltage 510 (i.e. shorter intervalsbetween times t1 and t2 and between times t3 and t4). As illustrated inFIG. 3, multiple charge transfer stages 340 may be provided.Functionality of the ADC 300 is enabled by the inclusion of at least asingle charge transfer stage 340. However, the inclusion and use ofmultiple charge transfer stages 340 may be used to increase the rampmaximum speed (i.e., the minimum interval required for a ramp signal togenerate all of the desired comparison values in the desired range ofcomparison voltages). The ramp maximum speed critical step is increaseat least in part by successively turning on transfer gates 334 inrespective charge transfer stages 340 to generate the ramp, as opposedto simultaneously turning on one or more transfer gates 334 to generatethe ramp. With successive charge transfers to the floating node 308 bymultiple charge transfer stages 340 being staggered in time (i.e., beingnon-simultaneous), the charge transfers from the supply 306 to thepinned photodiode devices PPD_1 in respective charge transfer stages 340may also be staggered in time. Specifically, the fill gates 332 inrespective charge transfer stages 340 may be successively activated tofill the pinned photodiodes PPD _1 to either their full well capacity orto any other capacity determined by the voltage provided at the supply306.

The inclusion of multiple charge transfer stages 340 also affords theADC 300 with built in redundancy that ensures operability of the ADC 300in the event that some of the PPDs PPD _1 in the charge transfer stages340 are faulty. Multiple charge transfer stages 340 also enable the rampstep size (i.e., the difference between V1 and V2, and between V2 and V3in the illustration of FIG. 5) to be varied, specifically by increasingthe number of charge transfer stages that transfer charges to thefloating Cdac1 node 308 at a given time. To compensate for full-wellcapacity variation in the pinned photodiodes PPD_1 in the chargetransfer stages 340, the ADC 300 may be operated to randomly select oneor more charge transfer stages 340 from the multiple charge transferstages 340 to generate each voltage step (i.e., each LSB for theconversion) to randomize noise caused by the pinned photodiode PPD_1full well variation, which may reduce gain mismatch between multiplecolumns of ADCs 300, when multiple ADCs 300 are implemented on an imagesensor.

One disadvantage of the architecture of FIG. 3 is the change in commonmode for the comparator in its switching point between performing theconversion for reset and signal. Specifically, the switching point ofthe comparator at a low voltage levels (such as when the pixel resetvalue is converted) may be different from the switching point of thecomparator at higher voltage levels (such as when the pixel signal valueis converted). Special design considerations are needed to make sure thecomparator does not generate an input offset that changes with thecommon mode level for the comparator switching point.

FIG. 6A illustrates a pixel signal level shifter 600 that whenincorporated into an ADC 700 (of FIG. 7), obviates concerns about thecomparator 318 of FIG. 3 having a first common mode switching level whenconverting a pixel reset charge value and having a second common modeswitching level when converting a pixel signal charge value. The ADC 700in FIG. 7 is provided the output 672 of the level shifter 600 at aninput 772 that is provided at the respective source-drain terminals ofthe charging transistor 704 and the comparison node pass transistor 782.Operation of the ADC 700 in FIG. 7 is similar to the operation of ADC300 of FIG. 3 described above, in that the conversion begins with thepixel reset charge being transferred to the Vref_comp capacitor 712 thatis coupled to the negative input of the comparator 718 via one of thecoupling capacitors 716.

However, in the FIG. 7 embodiment of ADC 700, the pixel reset chargethat is transferred to the Vref_comp capacitor 712 has been inverted andlevel-shifted by level shifter 600 of FIG. 6A. An illustrative range ofvoltages that may be present at the input 616 of the level shifter 600may be 1.5 V (for a pixel reset charge level) to 0.7 V (for a pixelsignal charge level). Generally, the level shifter 600 in FIG. 6A may beused to invert and shift the input voltage range of 1.5 V to 0.7 V to anoutput voltage range of 1.5V to 2.3V. Specifically, in response toreceiving a voltage of 1.5V at the input 616, level shifter 600 mayoutput a voltage of 1.5 V at the output 672 of level shifter 600. As thevoltage at the input 616 decreases to 0.7 V, the voltage at the output672 increases to 2.3 V. In other words, as the voltage at the input 616decreases within a first range of input voltages, the voltage at theoutput 672 increases within a second range of output voltages.

The first and second ranges may be determined at least in part by thecontrol signal provided to the variable capacitor 662 in level shifter600. The variable capacitor 662 may be connected to a negative terminal668 of an op-amp 618, which receives a common mode voltage Vcm at apositive terminal 670. The common mode voltage Vcm provided at thepositive terminal 670 of op-amp 618 may determine, at least in part, therange of output voltages produced at the output 672 of level shifter600. The level shifted pixel voltages stored on the floating Cdac1 node708 must stay above the pin voltage Vpin of the PPD_1 devices in thecharge transfer stages 740 (which, as an example may be 1.5 V) in orderto enable complete charge transfer from the pinned photodiodes PPD_1 tothe floating Cdac1 node 708. For additional margin in ADC 700, aslightly higher common mode voltage Vcm may be provided to the levelshifter 600, (such as 1.7 V, when Vpin is 1.5 V) to make sure theconversion of the reset value or very low signal values does not dropbelow Vpin (or, 1.5V in this example).

Between sampling operations of different pixels, the clamping switches714 are enabled (i.e., to create a connection between the Vclamp supplyand respective terminals of the coupling capacitors 716) as shown by theassertion of the auto-zero signal in FIG. 7B at time t3. After releasingthe clamping switches 714, the pixel reset level may be transferred tofloating Cdac1 node 708 and the Vref comp comparison node 784 byasserting the transistors 704 and 782, respectively (at time t3 of FIG.7B). Notably, the pixel reset level that is transferred to the floatingCdac1 node 708 and the Vref comp comparison node 784 is an inverted andflipped voltage that is produced at the output 672 of the level shifter600, indicated by the voltage level 1.5V at the Pixel level shifted inFIG. 7B using the exemplary values of a pixel reset level of 0.7V. Theconversion of the pixel reset level to a digital value is performed bythe successive operation of at least one charge transfer stage 740(represented as interleaved charge transfers 1940 and 1942) to thefloating Cdac1 node 708 to generate a decreasing ramp signal 610 that ina manner similar to the method described in connection with FIG. 5. Thecharge transfers 1940 and 1942 need not be interleaved, but can insteadbe distributed in time in any other suitable manner. Because the pixelreset level is provided at both the floating Cdac1 node 708 and theVref_comp comparison node 784, the counter 720 may be enabled for only afew clock cycles, during which at least one of the charge transferstages 740 may be used to fill their at least one respective pinnedphotodiodes PPD_1 and synchronously transfer the charges from the atleast one pinned photodiode PPD_1 to the floating Cdac1 node 708 tolower the voltage across Cdac1 capacitor 710, before being disabled atevent 1904 of FIG. 7B when the voltage at the positive input ofcomparator 718 is exceeded by the voltage at the negative input ofcomparator (i.e., the voltage across the Vref_comp capacitor 712).

To sample and convert the pixel signal level, after the pixel resetlevel has been converted, the transistor 704 is enabled in ADC 700 attime t12 with the assertion of the SHS signal to transfer the pixelsignal level (that has been level shifted by level shifter 600subsequent to the transfer of the pixel signal at t11 by level shifter600 when the Pixel transfer is asserted) to the floating Cdac1 node 708.In ADC 700, the coupling capacitors 716 are not clamped byenabling/closing the switches 716, after the pixel signal levelconversion is performed (i.e., the auto zero signal is not enabled aftert11 during the conversion process of the pixel signal level). Instead,the comparison node 784 maintains the pixel reset level across the Vrefcomp capacitor 712, while only the floating Cdac1 node 708 receives thepixel signal level output by the level shifter 600.

Once the pixel signal level has been transferred to the floating Cdac1node 708 , the conversion of the pixel signal level may commence withthe generation of a ramp voltage 610 illustrated in FIG. 6B.Specifically, the initial pixel signal level (such as V1 in FIG. 6B, forexample) may be decreased by an amount based on the full-well orpartial-well capacities of at least one pinned photodiode in at leastone respective charge transfer stage 740, and also based on the size ofthe Cdac1 capacitor 710, to a voltage level V2.

Charge transfers from charge transfer stages 740 may occur synchronously(i.e., every clock cycle) and may decrease the voltage across the Cdac1capacitor 710 by a fixed step size corresponding to the LSB valuerepresented by the converted digital value. Charge transfers may also beinterleaved as is illustrated in FIG. 7B where charge transfers 1944 and1946 are interleaved to create the decreasing voltages at the Vdac1 node708, which in turn effects a proportional change (specifically,decreasing voltage ramp) at the Comparator_VIN_P node. After everycharge transfer from stages 740 (i.e., at every clock cycle, or afterevery charge transfer operation 1944/1946), a count value stored in thecounter 720 may be incremented, provided that the voltage across thereference capacitor 712 (i.e., the voltage corresponding to the pixelreset level) is exceeded by the voltage across the Cdac1 capacitor 710after the charges have been transferred by the at least one chargetransfer stage 740 used to transfer charge to the floating node 708. Thecount value stored in the counter 720 is output when the voltage acrossthe Cdac1 capacitor 710 is exceeded by the voltage across the Vref compcapacitor 712 (i.e., just as the ramp voltage 610 at node 708 decreasesbeyond the reset level held at node 784) as is illustrated by the event1906 of FIG. 7B. In this way, the count value maintained by the counter720 is proportional to the number of voltage steps (or, charge transfersfrom at least one charge transfer stage 740) that are required to lowerthe pixel signal voltage to the pixel reset voltage, which is in turnproportional to the magnitude of the pixel signal voltage.

As an example, when a given pixel is operated in low-light conditions,the pixel signal provided at the input 616 of the level shifter 600 maybe close to or slightly below 1.5 V (i.e., close to or below the pixelreset level of 1.5 V); consequently as described above, the voltageproduced at the output 672 may be close to or slightly above 1.5 V.Because the voltage close to 1.5 V at the output 672 is provided at thesource-drain terminal of transistor 704 and then asserted at thefloating node 708 when the gate of transistor 704 is activated, thenumber of charge transfer operations to floating node 708 by stages 740before the voltage across the Cdac1 capacitor 710 is exceeded by thevoltage across the Vref_comp capacitor 712 (i.e., the pixel reset level612 output by the level shifter 600) may be small. When only a smallnumber of synchronous charge transfers are required to reduce thevoltage across the Cdac1 capacitor 710 below the voltage across theVref_comp 712, the count value maintained in the counter 720 may also besmall.

When a given pixel is operated in bright-light conditions, the pixelsignal provided at the input 616 of the level shifter 600 may be closeto or slightly above 0.7 V; consequently as described above, the voltageproduced at the output 672 may be close to or slightly below 2.3 V.Because the voltage close to 2.3 V at the output 672 is provided at thesource-drain terminal of transistor 704 and then asserted at thefloating node 708 when the gate of transistor 704 is activated, thenumber of charge transfer operations to floating node 708 by stages 740before the voltage across the Cdac1 capacitor 710 is exceeded by thevoltage across the Vref comp capacitor 712 (i.e., the pixel reset level612 output by the level shifter 600) may be large. When only a largenumber of synchronous charge transfers are required to reduce thevoltage across the Cdac1 capacitor 710 below the voltage across the Vrefcomp 712, the count value maintained in the counter 720 may also belarge.

Returning to the issue of the comparator 318 in ADC 300 potentiallyhaving different common mode switching values depending on whether apixel reset or a pixel signal level is being converted, the comparator718 in ADC 700 does not have any such issues. Because the voltage at thecomparison node 784 is constant for both the conversion of the pixelreset level and the pixel signal level, the switching point (or, thevoltage below which one of the inputs to the comparator 718 must drop,to flip or switch the output value of the comparator 718) of thecomparator 718 may be relatively constant. The switching point constancyis enabled at least because the value across the Vref comp capacitor 712may be constant for both the conversion of the pixel reset level and thepixel signal level, as the clamping switches 714 are not activated andthe transfer transistor 782 is not reasserted after the pixel resetlevel that has been shifted by the level shifter 600 has been assertedat the comparison node 784.

Comparators such as comparators 318/718 may be provided offsets tocalibrate the switching behavior at a given signal level. However, theswitching behavior of the comparator is not well defined (due to noise,at least) when the comparator is switched at another signal level thatis different than the given signal level at which the comparator 318/718was calibrated with offsets.

Charge transfer circuits such as 340 and 740 in FIGS. 3 and 7,respectively, may be used for implementing asuccessive-approximation-register (SAR) ADC 800, illustrated in FIG. 8A.FIG. 8C illustrates a graph of the voltages across the capacitors 810and 812 in an exemplary operation of the ADC 800. The SAR ADC 800includes two sets of charge transfer circuits 840 and 850. The chargetransfer circuits 840 and 850 may include n-type photodiode wells intheir respective pinned photodiodes PPD _1 that are used to transferpackets of negative charges (i.e., electron packets) to the Vdac1 node808 and the Vdac2 node 809, respectively.

The pixel signal provided at the input 872 may be a pixel signal (orpixel reset level) directly read out from an image pixel, but may alsobe a level shifted pixel signal (or level shifted pixel reset level). Asimplified level shifter 801 may also be used to provide the level shiftto the pixel signal or pixel reset level. Level shifter 801 of FIG. 8Bmay be used to shift the signal 816 produced by an image pixel. Thesignal 816 may be a pixel signal level or a pixel reset level. A pixelvalue 816 (i.e., the output of an image pixel, or “Pixel out”) may beprovided at an input 816 coupled to a positive input of an amplifier 878of the level shifter 801. A fixed voltage source 872 may be coupled at anegative input of amplifier 878. The fixed voltage source 872 may becoupled between the negative input of amplifier 878 and the output 872of amplifier 878.

The level shifter 801 of FIG. 8B may be used to apply a fixed voltageoffset, such as 0.5 Volts, 1 Volt, 1.5 Volts, or any other voltageoffset, to the image pixel output 816 provided at the positive input ofamplifier 878. In the example of FIG. 8B, the level-shifted signalproduced at the output 872 of amplifier 878 may be offset from the imagepixel output 816 by 1 Volt. In other words, for an input 816 to levelshifter 801 of 0.7 Volts, the output 872 of level shifter 801 may be 1.7Volts; for an input 816 to level shifter 801 of 1.5 Volts, the output872 of level shifter 801 may be 2.5 Volts. In this way, even when afully saturated pixel signal level is provided at the input 816 to thelevel shifter 801 (i.e., when the input 816 is provided a low voltage),the level-shifted version of the pixel signal level may be sufficientlyabove the pinning potential applied to the pinned photodiodes PPD _1 incharge transfer stages 840 and 850 (assumed to be at a pinning potentiallevel of 1.5 Volts, for illustrative purposes).

The size of the charge packets transferred by any one of the chargetransfer stages 840 or 850 may be determined at least in part by thevoltage provided at ground supplies 806 or 838, respectively. The sizeof capacitors Cdac1 810 and Cdac2 812 at the Vdac1 node 808 and theVdac2 node 809 respectively may determine, at least in part, the changein voltage at the nodes 808 and 809 that results from a charge packetfrom the charge transfer stages being transferred to one of the nodes.For a fixed packet size, a smaller capacitor Cdac1 810 may increase thechange in voltage produced at the Vdac1 node 808. Similarly, a largercapacitor Cdac1 810 may decrease the change in voltage produced at theVdac1 node 808. The resolution of the ADC 800 (i.e., whether ADC 800 isa 10-bit, 12-bit, or any other bit-resolution ADC) within a givenvoltage range of values that can be converted by ADC 800 may be adjustedby varying the voltages provided at ground supplies 806 and 838, and/orby varying the capacitances of the capacitors 810 and 812. The number ofSAR latches 880 and the amount of signals in the control bits 882provided to the SAR latches 880 may be based at least in part on thebit-resolution of the ADC 800.

Operation of ADC 800 may commence when a level shifted pixelsignal/reset voltage is provided at the output 872 of the level shifter801. A transfer transistor 882 may be used to pass the pixel voltagefrom the output 872 of level shifter 801 to the Vdac2 node 809. TheVdac2 node 809 may be coupled to a Cdac2 capacitor 812. The voltage atVdac2 809 may be the voltage across the Cdac2 812 capacitor. Assertingthe gate of the transfer transistor 882 may charge the Vdac2 node to thelevel shifted pixel voltage level. While the Vdac2 node 809 is chargedto the level shifted pixel voltage level at the amplifier 878 output872, the Vdac1 node 808 may be charged to the voltage level at pixelsupply voltage terminal 802 by asserting the gate of the pre-chargetransistor 804. The method of operating ADC 800 may be detailed in thetiming diagram of FIG. 8D. Between time t1 and t4, the pixel may bereset (when the pixel reset signal is asserted), and the pixel resetlevel may be sampled onto the Cdac2 capacitor 812 between time t3 andtime t5 at the Vdac2 node 809 (when the SHR signal is asserted).

A precharge transistor 802 may be coupled between the supply terminal802 and the Vdac1 node 808, and may be used to charge the Vdac1 node 808to a pixel supply voltage VAA at time t2, when the pre-charge transistor804 gate signal is deasserted to turn on the pre-charge transistor 804,prior to the conversion of the shifted pixel reset level provided atinput 872. As shown on the Pixel_output and Pixel level shifted lines ofFIG. 8D, when at time t1 the pixel reset is asserted, the Pixel_outputrises to a voltage that is approximately 1.5 V, namely to the resetlevel of the pixel. The Pixel level shifted line may shift the pixelreset level that is close to the 1.5 V by 1 V, to 2.5 V, and provide theshifted voltage to the input 972 to the sampling transistor 982.

At time t3, the precharge transistor may be activated to charge theVdac1 node 808 to the pixel supply voltage level. Consequently, thevoltage level at the Vinp node that is coupled to the Vdac1 node 808rises to the pixel supply voltage level. For simplicity of explanation,assume the Vdac1 808 voltage swing is shifted down from 2.8V-2V to2.5V-1.7V, to match approximately the range of the Vdac2 node whichranges from 2.5V to 1.7V. This offset at the Vdac1 node 808 may beimplemented by charge transfers from charge transfer circuits 840. Aswith the single slope ADC in FIG. 3, the ADC LSB size is set by the DACcapacitor size and PPD _1 charge capacity. In an example to illustratethe operation of ADC 800, the LSB is 0.8 mV and the full scale Vref ofthe ADC is 878 mV.

After the Vdac1 node 808 and the Vdac2 node 809 have been charged to thelevels provided at pixel supply voltage terminal 802 and the levelshifted output 872 of amplifier 878, the comparator 818 may compare thevoltages across the capacitors Cdac1 810 and Cdac2 812). The voltagesacross capacitors Cdac1 810 and Cdac2 812 may be referred to as thevoltages at Vdac1 808 and Vdac2 809, respectively. The pre-charge levelof Vdac1 is shown as V1 in FIG. 9. Prior to time t1, the pre-chargelevel V1 may be established at Vdac1 808 and the pixel voltage level V2may be established at Vdac2 809. Prior to time t1, the comparator 818output may be 1, indicating a higher voltage level at Vdac1 808 relativeto the voltage level at Vdac2 809.

The Vinn node that is coupled to the Vdac2 node 809 may be at the pixelreset level after the SHR signal is asserted at time t3. At time t6, thefill voltage Vfill1, corresponding to the voltage provided at the fillsupply terminal 806 may be dropped from 2.8 V to 0 V in the transition2052. However, the voltage levels that are used for the transition 2052are merely illustrative. The Vfill1 and Vfill2 voltages (the latterrepresenting the voltage at the supply 838 for the charge transfercircuits 850), may be adjusted so that the amount of charge in eachcharge fill operation of the charge transfer stages 840 or 850 may beadjusted. At time t7, the charge transfer stages coupled to the Vdac1node 808 may commence charge transfer operations 2040 and 2042. In theexample of FIG. 8D, 64 of the charge transfer stages (i.e., stages 840-1to 840-N, when N is 64) may be used to transfer charges to the Cdac1capacitor 810 at time t7.

The transfer of charges may proceed as described in connection with thecharge transfers 1840 of FIG. 3. SAR latch values may be determined forthe pixel reset levels according to the flow chart of FIG. 8E. At step2090, a first number X of PPD devices in charge transfer stages 840 maybe transferred a second number Y amount of times to the Cdac1 capacitor810. The product of X and Y may be the net number of charge transfersthat occur in step 2090. At step 2092, the voltage at the Vdac1 node 808may be compared to the voltage at the Vdac2 node 809 by the comparator818.

If, as a result of the comparison, it is determined that the Vdac1 node808 has a voltage level higher than the voltage level at the Vdac2 node809, step 2094 may be performed. In step 2094, the N-th bit (i.e., theMSB, for the first iteration of the method in FIG. 8E) may be set to 1.N may then be decremented so that subsequent iterations of the method ofFIG. 8E set the N-1-st bit. Finally, the net number of charge transfersmay be adjusted by adjusting either the number X of PPD devices incharge transfer stages 840 that are to be subsequently filled, or thenumber of times Y that said PPD devices in charge transfer stages 840are to be billed, or both X and Y. As an example, if X was 128 and Y was4 at step 2090 prior to step 2094, at step 2094, X may be changed to 64,while Y is kept as 4. X may alternatively be kept at 128 while Y ischanged to 2, or X may be reduced to 32, while Y is raised to 4.Generally, X and Y may be chosen such that the net number of chargetransfers in immediately prior instances of step 2090 (or immediatelyprior instances of step 2098) is halved. After step 2094, step 2090 maybe performed again.

If, however, as a result of the comparison it is determined that theVdac1 node 808 has a voltage level lower than the voltage level at theVdac2 node 809, step 2096 may be performed. In step 2096, the N-th bit(i.e., the MSB, for the first iteration of the method in FIG. 8E) may beset to 0. N may then be decremented so that subsequent iterations of themethod of FIG. 8E set the N-1-st bit. Finally, the net number of chargetransfers may be adjusted by adjusting either the number X of PPDdevices in charge transfer stages that are to be subsequently filled, orthe number of times Y that said PPD devices in charge transfer stagesare to be billed, or both X and Y, though these numbers will relate tothe number and operations of charge transfer stages 850 that are coupledto the Vdac2 node 809. Similar to step 2094, X and Y may be chosen suchthat the net number of charge transfers in immediately prior instancesof step 2090 (or in immediately prior instances of step 2098) is halved.After step 2096, step 2098 may be performed, in which a number X ofcharge transfer stages 850 are used to transfer charges to Vdac2 809 Ynumber of times. After step 2098, step 2092 may be performed.

From time t5 to time t10, the pixel reset level may be converted. Chargetransfer operations 2040 and 2042 from the charge transfer stages 840may occur during this interval to transfer charges to the Vdac1 node 808(i.e., when step 2090 of FIG. 8E occurs), and charge transfer operations2044 and 2046 may occur during this interval to transfer charges to theVdac2 node 809 (i.e., when step 2098 of FIG. 8E occurs). The Vfill1 andVfill2 voltages for the supplies 806 and 838 respectively, may beadjusted simultaneously. As an example, the transitions 2054 and 2058,2056 and 2060, and 2062 and 2064 may be coordinated. At saidtransitions, the Vfill1 and Vfill2 voltages for the supplies 806 and 838may be switched from a low voltage level to a high voltage level, orvice versa. At time t8, the pixel signal level may be transferred to afloating diffusion node of the pixel. The pixel output (“Pixel_output”)may shift by an amount Delta V1. The shifted pixel output (i.e.,“Pixel_level_shifted” output from the level shifter 801) may also shiftby an amount Delta V1 to a level that is 1 V (using the exemplary valueused for description) higher than the level of the pixel output. At timet10, the pixel row select may be deasserted. At time t11, the prechargecontrol voltage to the precharge transistor 804 may be deasserted,thereby precharging the Vdac1 node to the pixel supply level VAA. Alsoat time t11, the SHS signal provided to sampling transistor 882 may beasserted, transferring the shifted pixel level provided at input 872 tothe Vdac2 node 809. From time t11 onward, the SAR conversion of thepixel signal level may proceed in the manner detailed in FIG. 8E.

An illustrative example of the method of FIG. 8E is presented below toclarify the operation of the SAR ADC 800. To determine themost-significant-bit (MSB) of the digital value representing the pixelvoltage level V2 at Vdac2 809, multiple charge transfer stages 840 maybe used to fill pinned photodiodes PPD_1 and transfer charges from thepinned photodiodes PPD_1 to the Cdac1 capacitor 810. In an illustrativeexample where at least 64 charge transfer stages 840 are provided in anADC 800 of FIG. 8A, 64 charge transfer stages 840 may be filled (i.e.,the pinned photodiodes PPD_1 in 64 charge transfer stages 840 may befilled by asserting the gates of the fill transistors coupled betweensaid pinned photodiodes and the supply terminal 806), and thentransferred/dumped to the Cdac1 capacitor 810 (i.e., after the pinnedphotodiodes PPD_1 in 64 charge transfer stages 840 are filled, thecharges in the pinned photodiodes PPD_1 may be transferred to the Cdac1810 capacitor by asserting transfer transistors coupled between saidpinned photodiodes and the Cdac1 capacitor 810). The 64 charge transferstages 840 may be filled and dumped to the Cdac1 capacitor 810 anadditional 7 times (for a total of 8 total transfers from the 64 chargetransfer stages 840) to lower the pre-charge voltage V1 to a voltage V3at time t1.

After the 64 charge transfer stages 840 have been used to transfer theirrespective charge packets to the Cdac1 capacitor 810 eight (8) times,the comparator 818 may compare the voltages at the Vdac1 node 808 andthe Vdac2 node 809.

In response to determining that the voltage at the Vdac1 node 808 isgreater than the voltage at the Vdac2 node 809, the comparator 818 mayoutput a logic high voltage level (i.e., a logic “1” voltage level) tothe SAR latches 880. In response to receiving a logic high voltage levelfrom the comparator 818 after the first charge dump at time t1, the SARlatches 880 may store a logic high voltage at a latch that represents aMSB of a multi-bit digital value or quantity.

In response to determining that the voltage at the Vdac2 node 809 isgreater than the voltage at the Vdac1 node 808, the comparator 818 mayoutput a logic low voltage level (i.e., a logic “0” voltage level) tothe SAR latches 880. In response to receiving a logic low voltage levelfrom the comparator 818 after the first charge dump at time t1, the SARlatches 880 may store a logic low voltage at the latch representing theMSB of the multi-bit digital value.

Voltage at the Vdac2 node 809 being greater than the voltage at theVdac1 node 808 at time t1 indicates that the charge dumped onto theVdac1 node 808 corresponding to an amount of charge represented by amulti-bit digital value with a logic “1” only at its most significantbit is greater than the pixel signal (i.e., the voltage at the Vdac2node 809 at time t1). In traditional SAR-based ADCs, a subsequentcomparison would involve producing a voltage representing anintermediate voltage between the first tested voltage (i.e., the voltageV3 at time t1, corresponding to a digital value with a logic “1” only atits most significant bit) and the voltage indicating a digital valuemade entirely of logic “0” values (i.e., the voltage V1). This oftenoccurs by adding a voltage to the previously generated voltage forcomparison to the pixel voltage level.

For SAR ADC 800 however, generating a voltage that is greater than V3 ata time subsequent to time t1 may not be possible, at least when usingthe charge transfer stages 840 to change the voltages at the Vdac1 node808. Because the charge transfer stages 840 and 850 transfer electronpackets to the Vdac1 808 and Vdac2 809 nodes respectively, the voltagelevels at nodes 808 and 809 may only be reduced by the charge transferstages 840 and 850, but not raised.

To enable SAR ADC conversion, however, the comparison of the pixelvoltage level to a voltage level greater than the current voltage muststill occur (at least when the voltage at the Vdac2 node 809 is greaterthan the voltage at the Vdac1 node 808 and the comparator 818 output isa logic low level or logic “0”). However, instead of increasing thevoltage at the Vdac1 node 808 by a given amount, the voltage at theVdac2 node 809 may be reduced by the given amount. One or more chargetransfers from one or more charge transfer circuits 850 coupled to theVdac2 809 node can used to decrease the voltage at the Vdac2 node.Charge transfer circuits 850 may decrease the voltage at the Vdac2 node809 by an amount required to determine whether or not thesecond-most-significant bit (sometimes referred to as MSB-1) should beset to be a logic high value at the latches 880.

To determine the MSB-1 bit in a corresponding one of the latches 880, 64of the charge transfer circuits 850 may be filled and dumped to theVdac2 809 node 4 times at time t2 (i.e., half of the number of chargetransfers compared to the charge dump at t1, from the same number ofcharge transfer circuits used in the charge dump at t1).

In the example shown in FIG. 9, between times t2 and t3, the Vdac1voltage level V3 is greater than Vdac2 level V4, and the MSB-1 bit inthe SAR latches 880 is accordingly set to “1.” Next, the bit MSB-2 isdetermined by shifting Vdac1 by ⅛ the ADC reference range to a voltageV4 at time t3. To shift Vdac1 by ⅛ the ADC reference range, 128 PPD _1charge packages from charge transfer circuits 840 may be transferred toCdac1. To achieve this, the entire bank of 64 PPD_1 may be filled anddumped to the Vdac1 808 node 2 times.

Note that the MSB-3 bit determination may require just 64 PPD_1transfers (i.e., a single transfer of charge packets from 64 chargetransfer circuits 840/850) at time t4. The subsequent bit determinationsrequire only a subset of the PPD bank to be transferred (i.e., only 32,16, 8, 4, 2, 1 charge transfer circuits 840/850 need to be used totransfer charge packets a single time).

The advantage of this SAR architecture using PPD charge packets is sizerelative to a binary scaled capacitor based approach (especially M-i-Mcapacitors) and this approach does not require accurate capacitormatching or capacitor voltage linearity. For advanced technology nodes,capacitor options are limited and voltage linearity of available MOScapacitors limits SAR bit depth.

As with the single slope ADC design of FIG. 3, the ADC reference of FIG.8A may be changed by modifying the fill voltage for the PPD or the sizeof the Cdac capacitor. Also, the number of charge transfer stages840/850 in the ADC 800 can be changed to make a faster or slower SARdepending on area constraints.

In order to reduce routing congestion in the circuit especially in acolumn parallel configuration, the “Fill” and “transfer” control signalthat respectively control the fill and transfer transistors in a chargetransfer stage 840/850 may be globally controlled for all columns andthe fill voltage for the bank of PPD can be controlled locally. Bycontrolling the PPD fill voltage (drain node of the “Fill” transistors),the local circuit can set how much charge is added to the PPD (zerocharge or fixed charge) while the global control signals can be set toenable transfer regardless of the internal ADC state (the state of thecomparison between Vdac1 808 and Vdac2 809 nodes determining the nodethat receives charge packets from the charge transfer circuits 840/850).

FIG. 10 illustrates an implementation 900 of the SAR ADC 800 with moredetails about the auto-zero 928 of the comparator 918 and clampingcapacitors 916 at the input of the comparator 918 to decouple the Vdac1908 voltage common mode from the Vdac2 809 common mode level. In the SARADC designs of FIGS. 8A and 8C, the comparator 818/918 must be designedto suppress any offsets from changes in common mode. Again, thisrequirement is because the SAR ADC 800/900 uses a differential topologywhere the comparator switching point changes depending on the signallevel being converted.

The clamp switches 914 and comparator auto-zero 928 (the latter of whichmay be omitted) are activated when the Cdac1 capacitor 910 ispre-charged to VAA (i.e., the supply 902 voltage level) and the pixellevel shifted reset voltage (provided by the output of the amplifier inthe level shifter 901) is sampled on the Cdac2 capacitor 912. Then, theSAR conversion may be performed. For the next SAR conversion (i.e.,after the conversion of the level-shifter pixel reset signal iscomplete) the Cdac1 capacitor 910 is pre-charged again to the supply 902level VAA and a pixel level shifted signal voltage is sampled on theCdac2 capacitor 912, without the clamps to the Cc capacitors beingactivated (similar to the single slope ADC sampling scheme).

Note that an inverting amplifier could be used as well (like used inFIG. 6A for the single slope ADC) as long as the ADC 800/900 logic isadapted to the change in signal and reset polarity.

Note that with timing changes, the SAR can operate during mostsignificant bit determination in a single sided mode (where only Cdac1910 changes voltage during conversion) as opposed to a double sided mode(where both Cdac1 910 and Cdac2 912 change voltage during conversion).This is possible by operating Cdac1 in an iterative fashion afterdetermining the first few most significant bits (2 or 3) and thensubsequently pre-charging Cdac1 910. After precharging Cdac1 910, thenthe charge transfer stages 950 bank may iteratively transfer charge toCdac1 910 according to how the MSBs are set. Then the remaining leastsignificant bits are determined in the double side mode where smallvoltages happen on both Cdac1 910 and Cdac2 912. The smaller voltagechanges on Cdac2 912 reduce the required operating input range for thecomparator 918.

For the iterative MSB determination described above, extra time isneeded to reload the ADC 900 with the MSB charge on Cdac1 910 andre-load the input on Cdac2 912. In order to speed up the first passdetermination of these bits, it is possible to change the value of Cdac1910 and Cdac2 912 to smaller value during the first pass in order torequire less charge packet transfers to them (e.g. ½ smaller capsrequire ½ the number of packet transfers). Then Cdac1 910 and Cdac2 912are returned to their final capacitance value for the remaining bits inorder to achieve the target ADC noise requirement.

An improvement on the design of FIG. 9 is shown in FIG. 10. The designof FIG. 10 does not require the level shifter 801/901 on the pixelinput. The original purpose of the level shifter 801/901 is to keep theCdac2 812/912 node at a high enough voltage (i.e., greater than thepinning potential of the pinned photodiodes PPD_1 in the charge transfercircuits 950) during operation to allow charge transfer from the pinnedphotodiodes PPD_1 in the charge transfer circuits 950 to Cdac2 912. Thatrequirement can also be met by shifting the voltage on the bottom ofCdac2 capacitor (1012 in FIG. 12) to a higher voltage during chargetransfer, using a voltage shifter 1088 that is coupled to the Cdac2capacitor 1012. After the voltage shifter 1088 has been used to shiftthe voltage at the bottom of the Cdac2 capacitor voltage up, the bottomof Cdac2 is return to 0V during other operations (sampling the pixelinput, clamping the comparator, or during comparator strobe).

A comparator topology 1100 that is not sensitive to common mode offsetsand uses the PPD charge transfer circuits is shown in FIG. 11. It isconnected to the SAR ADC 1101 on the left (i.e., the components1102-1112 1140, 1150, 1172, 1182, and 1188 may be substantiallyidentical to similarly numbered components in FIGS. 9 and 10). Thecomparator 1118 operates by sampling charge from both Cdac1 1110 andCdac2 1112 into PPDs (i.e., PPD_sample pinned photodiodes in thepseudo-pixel circuits 1150). The charge from these sampling PPDs istransferred to floating diffusion 1162 structures as used in a pixel andthe source followers 1158 are configured together to determine which ofthe capacitors of Cdac1 1110 and Cdac2 1112 has the higher voltage.

In order to sense the charge on Cdac1 1110 and Cdac2 1112, they arefirst level shifted down using the voltage shifter 1188 so that themaximum voltage on either capacitor 1110 or 1112 is 1.5V. This levelshifter 1188 is achieved with a standard coupling technique by changingthe voltage on the bottom of the Cdac capacitors 1110 and 1112 as shownin FIG. 11. With Cdac1 and Cdac2 voltages below 1.5V, both can samplecharge on the PPD devices labeled PPD_sample by asserting the gate ofsample transistors 1152 in the pseudo-pixels 1150. The capacitor withthe highest voltage will transfer the least number of electrons to therespective PPD_sample device. Then, the floating diffusions 1162 labeledvinp and vinn are reset to VAAPIX (2.8V). Then, the charge istransferred to vinp and vinn from the respective PPD_sample device byasserting the gate of a transfer transistor 1154 in the respectivepseudo-pixel 1150.

Note that the source follower 1158 drains are connected to separateoutput lines that are pre-charged to 2.8V (VAAPIX) by pre-chargetransistors 1176 and 1178, and the sources are tied together to acurrent source 1190. After charge is transferred to vinp and vinn, thepre-charge signal (i.e., the signal provided to the gates of pre-chargetransistors 1176 and 1178) is disabled and the source follower 1158 withthe highest gate voltage (i.e., the lowest number of charges at therespective floating diffusion 1162) will discharge the line to 0V. Thesource follower 1158 with the lower gate voltage will be off because itssource voltage is set high enough to turn off the source follower 1158and keep its output set to the high pre-charge voltage. Some capacitanceon the drains of the source follower 1158 is needed to create moremargin in the circuit between the “off” source follower and “on” sourcefollower.

Ideally, the capacitance Cfd 1156 is kept very low to maximize voltageresolution of the comparator 1118 (i.e., a capacitance that changesvoltage by less than 200 uV per electron). Also, it is important tominimize the signal sampled in the PPD_sample device because thedifference in charge sampled from Cdac1 and Cdac2 creates an errorsignal in the conversion. If 10 comparisons are performed per conversionand the maximum error accumulated during the conversion is ½ LSB (eachLSB equal to PPD_1 full capacity or 5000 electrons in this example),then the PPD_sample should only sample (5000/2)/10 electrons or 250electrons. Other tradeoffs like PPD_1 size or charge used to set eachDAC step can be maximized to reduce error from the comparator sampleoperation removing charge.

If this comparator design replaces the FD node with a “floating gate”NMOS transistor 1264, as shown in the comparator 1218 of FIG. 12, thenthe charge sampled into the circuit can be returned to the Cdac1 1110and the Cdac2 1112 capacitors. The floating gate of transistor 1264 isinitially set to a high voltage VAAPIX (2.8V) by the asserting the gateof the reset transistor 1260. It does not contain charge in its channel.Then charge is transferred to the channel with the transfer transistor1254 gate pulsed high and the circuit behaves like the operation abovewith the floating gate of transistor 1264 driving the source follower1260 gate (any charge transferred under the floating gate will pull thesource follower gate voltage down). At the end of the comparison, thetransfer gate 1254 is turned on and the floating gate of the transistor1264 is set to 0V. Then, the channel charge is transferred back to thePPD_sample in the pseudo-pixel 1250 device. Cdac1 and Cdac2 bottomplates are returned to 1.2V to level shift them back to a highervoltage. Then, the gate of the sample transistor 1252 is turned on totransfer the PPD_sample charge back to the Cdac nodes 1110 and 1112.With this operation, the comparison is non-destructive and does notcreate an error charge on the Cdac nodes 1110 and 1112.

The charge summing technique described in connection with the aboveembodiments which use charge transfer circuits with pinned photodiodesto transfer/dump the electron packets onto a node is also useful forbuilding signal integrators as well. To enable a delta sigma ADC, forexample, both an integrator and a decimator/subtractor are needed torespectively sum and subtract signals. A pinned photodiode thataccumulates holes is needed to subtract signal (i.e., remove electrons)from an integrator summing node.

FIG. 13A shows a first order delta sigma modulator 1300 that uses acharge transfer circuits on the left bank 1340 of charge transfercircuits to inject electrons charge packets onto the summing node 1308and that uses a right bank 1350 of charge transfer circuits with holebased pinned photodiodes (labeled “HPD_1”) to inject holes onto thesumming node 1308. Notably, PMOS transistors may be used for the filland transfer transistors 1332-1 and 1334-1 connected to the hole basedpinned photodiodes in the right bank of charge transfer circuits. Inorder to fill a HPD with holes, a voltage of 3.5V is connected to thesource of the fill transistors such as 1132-1 and the gate of the filltransistor 1132-1 is asserted (i.e., the gate of the fill transistor1132-1 is biased with a logic “low” level voltage, given that the filltransistor 1132-1 is a PMOS transistor). After a HPD is filled withholes, the holes (now referred to as a “hole charge packet”) may betransferred to the summing node Vdac1 1308 when the DAC operation isneeded. Complete charge transfer of holes only happens when the DACsumming node 1308 is less than 2.5V because the pin potential of the HPDis −1V.

The higher 3.5V is shown for illustration purposes only. The Vpin forthe electron based PPD in the left bank 1340 and hole based HPD in theright bank 1350 may be adjusted to smaller magnitudes to enableoperation between 0V and 2.8V only. For example if the magnitude of bothVpins is 1.0V, then the summing node can operate between 1.8V(VAAPIX-1V) and 1.0V where complete charge transfer is possible(electrons from PPD and holes from HPD) for correct operation.

Notably, the HPD_1 device design is not needed for imaging and is onlyneeded to store and transfer holes. A hole based pinned photodiode maybe fabricated in an Nwell (tied to high voltage) with an analogous n+surface pinning layer and p implanted pinned photodiode. These circuitscan also be designed to be tolerant to some PPD lag from partial chargetransfer.

Gain of the integrator is set by the number of the PPD_1 transfers(i.e., electron charge packet transfers) from the charge transfercircuits in the left bank 1340 of charge transfer circuits. The pixelbuffer (or inverting amplifier) 1301 generates a level shift of thepixel output Pixel_out to a range from 1.5V to 0.0V where the PPD_1devices will fill up with electrons in proportion to the pixel outputlevel (ideally the high voltage output from the pixel is equal to Vpin).The DAC level subtracted from the integrator summing node isproportional to the number of HPD_1 transfers on the right side of thesumming node. The gain of the integrator is also set by the Cdac1 size.The amount of gain in the input signal path and DAC path is determinedby the conventional system design of this circuit.

The front end of the integrator can also be used to sum pixel values asthey write their values into the PPD_1 devices. This summing operationcan also use different weights for the pixel values summed bycontrolling the number of PPD_1 transfers.

The delta sigma modulator of FIG. 13A may be modeled using the blockdiagram of FIG. 13B. The pixel buffer 1301 that provides the inputvoltage may be represented by the gain block 1390 that applies a gain tothe Vin signal (or, a level shift, in the case of the circuitry of FIG.13A). The comparator 1318 may receive the Vref voltage (shown in FIG.13B as Vcomp) and the summing node value Vsum (corresponding to thevoltage at the Vdac1 node 1308 of FIG. 13A. The DAC 1352 may correspondto the hole based pinned photodiode charge transfer circuits in theright bank 1350 of FIG. 13A. The integrator 1342 may correspond to theelectron based pinned photodiode charge transfer circuits in the leftbank 1340 of FIG. 13A. The gain block 1394 may represent a gain that isprovided to the DAC 1352 output. The node 1392 may show that the DACvalue is subtracted, while the Vin input value is added (after beingscaled by their respective gain blocks 1394 and 1390) before beingprovided to the integrator 1342.

The first order delta sigma modulator of FIG. 13B is shown with gainblocks 1390 and 1394 in the input path (xG0) and feedback path (xG1) forgenerality. As an example, gains for both paths may simply be 1. Theintegrator sums the input voltage value until the comparator thresholdis reached and then subtracts off the ADC voltage reference value. Afterthe ADC reference value is subtracted, the integrator continues to sumthe input voltage until the comparator threshold is reached again.Typically, a digital counter or digital summing circuit counts thenumber of times the comparator threshold is reached. This counter isviewed as simple digital low pass filter that is a “box filter” becauseeach output equally increases the count value by 1 as opposed toweighting the filter input values to achieve a different frequencyresponse. With the addition of the DAC feedback, the average integratoroutput voltage over time (modulated by the input voltage and DAC voltagesignals) will approach the same value as Vin (or proportional to Vin)and the number of times the comparator threshold is reached relative tothe total number of summing operations is representative of the digitalvalue. In fact, the ratio of the total number of comparator 1'sgenerated to the total number of comparison operations multiplied by theADC reference voltage approximately equals the input signal voltage (orproportional to Vin). The high integrator gain at low frequencies andlarger total number of samples increases the accuracy of the modulator.

The comparator output can also be viewed as a 1-bit ADC that isconverting the integrator output. The “0” output represents a signalvalue of 0V and the “1” represents a signal value of Vref (full scaleADC response). Because the signal is between 0V and Vref, the ADC-DACcombination is feeding back to the integrator the quantization error(difference between ADC digital representation and the analog inputvalue). Because the modulator is clocked (for each summing operation) ata frequency much higher than the signal (set by the over samplingratio), this quantization error is being created at higher frequenciesthan the input signal and is being partially low pass filtered by theintegrator. While the integrator output voltage average converges to theinput signal (the combination of the integrator and feedback produceshigh signal gain at low frequencies that results in the integratoroutput average equaling the input signal), the integrator output hashigher frequency quantization noise superimposed on it that is notfiltered by the integrator. The digital filter following the comparatorremoves some of this high frequency noise by performing a low passfilter operation on the stream of 1's and 0's from the comparator. Thus,the strategy used is to create a sequence of digital values from thecomparator that represents the input signal level, push the quantizationnoise to higher frequencies outside of the signal bandwidth of interestand subsequently filter out the higher frequency noise content. For theADC timing diagram of FIG. 13C the Cdac1 node is first reset to 2.5V (asan example) with the deassertion of the reset signal to the resettransistor coupled between the 2.5V supply voltage and the summing nodeVdac1 1308 in the integrator 1340, from time t1 to time t2. Then theinput signal is sampled into the PPD _1 circuit with the Fill1/TX1 toFill64/TX64 bank of signals (e.g., the charge transfers 1388 and 1389,while the charge transfers from the remaining charge transfer stages inthe integrator 1340 are omitted from the illustration of FIG. 13C forsimplicity). For example, for a Cdac1 value of 0.2 pF and input voltagelevel that generates in the PPD 5000 electrons, a signal level drop of256 mV is generated on Cdac1 for 64 input samples taken at once. Thevoltage on the output of the integrator Cdac1 is shown at the bottom ofthe timing figure. After the input signal is sampled into the integratorCdac1, the comparator is strobed to see if the input is smaller than thecomparator reference set at 1.8V. If the input is smaller than 1.8V,then positive charge is added to the Cdac1 node (via charge transfers1382 and 1383 from hole based pinned photodiodes, for example) toincrease the voltage on the Vdac1 node 1308. For example, multipletransfers of the hole based HPD are used to transfer enough charge tomove the Cdac voltage by 0.7V that represents the ADC reference.

The input signal is sampled into the integrator again with thesubsequent drop in output of another 256 mV and the comparator isstrobed once again. This cycle continues until the comparator thresholdis crossed and in the next input sample phase, positive charge is addedto Cdac1 to shift the voltage by the ADC reference voltage that equals0.7V in this example. To sample positive charge to Cdac1, the signalsFill1h/TX1h to Fill64h/TX64h are pulsed. It is assumed that the hole HPDholds about 13,600 units of positive charge (holes) for this exampleonly to generate the 0.7V shift for 64 hole based HPDs charge transfer.This cycle of input sampling and comparison continues until theoversampling ratio is achieved (e.g. 1024 clocks) for the target ADCresolution.

An implementation of the first order delta sigma modulator 1300 thatdoes not require hole based PPD devices is shown in FIG. 14. A secondset of PPD_2 device stages 1452 with a higher photodiode pinningpotential of 2V is connected to the integrator 1340. When the integratorfills up with charge, the comparator flips (i.e., the value produced atcomparator output 1318-3 may change). At this point, the Skim_TX voltageprovided at the gates of the skimming transistors 1454 is set to a value(2.8V) that creates a barrier (1.8V) between the Cdac capacitor 1310 andPPD_2 that allows excess electrons beyond a certain charge capacity toflow into the PPD_2 devices in stages 1452. The Skim_TX transistor 1454may be turned off again and the electrons are then removed from theintegrator by precharging it again (i.e., the precharge transistor 1404may be activated). After precharge is complete, the charge in PPD_2 istransferred back to the Cdac capacitor 1310 by setting the Skim_TX gateof the skimming transistors 1454 to a high voltage (>3.0V) that does notcreate a barrier. Hence, the DAC value is subtracted from the integratorsignal and only the residue charge (from PPD_2) is remaining in theintegrator as with normal delta sigma modulator operation. Integrationcontinues until the next comparator flip.

FIG. 15 shows an implementation of a first order delta sigma modulator1500 that does not require a second Vpin potential. It shifts the bottomplate of the Cdac1 capacitor to a higher voltage using a voltage levelshifter 1588 to enable charge transfer from the PPD_1 devices in theintegrator 1340. Then, the bottom plate is returned to its lower voltagefor the comparison operation. If the comparator 1318 flips, then theSkim_TX voltage at the gates of the skimming transistors 1554 in PPD_2device stages 1552 is set to a value (2.3V) that allows excess charge toflow into PPD_2 that has the same pin potential (Vpin) as PPD_1 (e.g.1.5V). After precharge is complete (i.e., after the reset transistor1404 is activated), the charge in PPD_2 is transferred back to the Cdacnode by setting the Skim_TX gate of the skimming transistors 1554 to ahigh voltage (i.e., a voltage greater than 2.5V).

It is also possible to set the Cdac1 1510 bottom plate to a thirdpossible voltage during the precharge operation and comparator operation(e.g. 0.5V). Then during the Skim_TX operation when charge is beingskimmed off or removed from Cdac1 1510, the Cdac1 1510 bottom plate isset to 0V. This option gives more design flexibility in setting thebarrier between Cdac1 1510 and PPD_2 as well as the effective ADCreference.

It is expected that the thermal energy of electrons will create someuncertainty in charge crossing the barrier and “skimmed” off the Cdac11510 node. However, this uncertainty in electrons skimmed off will beaveraged out over time as multiple cycles are done in the delta sigmamodulator.

In order to increase the accuracy of the A/D operation, the overall lowfrequency gain of the circuit that pushes the quantization noise tohigher frequencies can be increased by cascading integrators to form asecond order sigma delta modulator as shown in FIGS. 16A and 16B. FIG.16B shows a block diagram view with additional gain is shown with thexG2 gain block 1696 and the xG3 gain block 1698. Part of the reason forthe different gain values is to prevent the integrator from moving outof range for operation of the circuits forming the integrator (also,these are set to control the magnitude of the frequency noise componentsof the quantization noise and the stability of the modulator). Bycascading modulators the quantization noise is also moved to higherfrequencies and the increased gain also allows the integrator outputsand fed back average comparator output to more accurately track theinput signal level. Along with the low pass digital filter, thistopology can increase the accuracy of the A/D conversion.

A second order delta sigma modulator 1600 is shown in FIG. 16. It ismade by replicating the first order modulator circuit. A first-ordermodulator circuit 1620-1 may include a first integrator 1640-1 and afirst subtractor 1650-1 that are both coupled to a summing capacitor1610-1. A first-order modulator circuit 1620-2 may include a secondintegrator 1640-2 and a second subtractor 1650-2 that are both coupledto a summing capacitor 1610-2. A unity gain buffer 1690 may be coupledbetween the modulators 1620-1 and 1620-2. Ideally, the unity gain buffer1690 shifts the level of the first integrator 1640-1 by exactly 1 V sothat the PPD _1 devices in the second integrator 1640-2 input fill upwith electrons in proportion to the first modulator 1640-1 output.Again, the gain coefficients of the two integrators input and feedbackpaths (i.e., the gain coefficients of integrators 1640-1 and 1640-2) isset by the number of transfers to each summing node from the inputs toeach stage and feedback DAC.

The second order delta sigma ADC timing diagram in FIG. 16C for FIG. 16Ashows the input sampling signals for the first (Fill1/TX1-Fill64/TX64,shown as charge transfers 1690 and 1691) and second integrator blocks(Fill1_m2/TX1_m2-Fill64_m2/TX64_m2, shown as charge transfers 1693 and1694). First, both modulators 1620-1 and 1620-2 Cdac nodes are reset to2.5V (i.e., the Cdac1 capacitors 1610-1 and 1610-2 are both reset to thesupply voltage by the deassertion of the reset signal, which is notshown in the timing diagram for simplicity). The first modulator addsthe input voltage from the pixel to the first integrator node (i.e., theCdac1 capacitor 1610-1) by means of charge transfers 1690 and 1691 fromtime t1 to time t2, for example. At the end of the first modulator inputsampling phase, the second modulator samples the first modulator outputby means of charge transfers 1693 and 1694 form time t2 to time t3, forexample. Then the comparator is strobed at the output of the secondmodulator from time t3 to time t4. The waveforms at the output of thefirst and second modulator output are shown in the figure as the valuesof the Cdac1 integrator node (for Cdac1 1610-1) and the Cdac2 integratornode (for Cdac1 1610-2). If the comparator output threshold is reached,positive charge is added to both first and second modulator Cdac nodesthrough charge transfers from the hole based pinned photodiodes. Theamount of charge added can be set by the number of hole based HPDsenabled to transfer charge. This amount can be different for the firstand second modulator.

The second order delta sigma ADC with improved signal range is shown inFIG. 16D. It allows the output of the first modulator 1620-1 to riseabove the Cdac reset level (e.g. 2.0V) to allow either positive ornegative charge to be added to the second modulator 1620-2. The topamplifier 1690-1 in the modulator works on input signals between 2.0Vand 1.5V to level shift them to 1.5V and 1.0V respectively. This levelshift puts the output of the amplifier in the range of the electronbased PPD devices to add electrons to the integrator. The bottomamplifier 1690-2 operates on input signals between 2.0V and 2.5V tolevel shift them to 2.5V and 3.0V respectively. This level shifts putsthe output of the amplifier in the range of the hole based HPD devicesto add holes to the integrator. The second modulator 1620-2 outputoperates in the range between 2.5V and 1.5V. The comparator has itsthreshold set at 1.8V. When the output of the second modulator dropsbelow 1.8V, positive charge is added to both first and second modulatorCdac nodes 1610-1 and 1610-2. The amount of charge added can be set bythe number of hole based PPDs enabled to transfer charge. This amountcan be different for the first and second modulator. This difference inthe amount of voltage added to the first and second order modulator isneeded to make sure the circuit stays within the operating range for theamplifier and integrators.

For example, xG0 and xG1 for the first modulator is equal to 0.3× wherethe voltage added to the integrator (through charge transfer) from theinput signal and Vref feedback path is multiplied by a factor of 0.3 bymodulating the number of times each is sampled into the integrator (alsodetermined by charge capacity of the PPD/HPD and Cdac size). Also, as anexample, xG2 and xG3 for the second modulator is equal to 0.7× where thevoltage added to the second integrator from the first modulator outputand Vref feedback path is multiplied by a factor of 0.7 by modulatingthe number of times each is sampled into the integrator.

The timing diagram of FIG. 16E for the improved signal range delta sigmaADC shows the second integrator output that can increase or decrease involtage based on the input level of the first integrator output level.Because of feedback in the circuit topology, the output of the secondintegrator will trigger the comparator so that the filtered sequence ofthe comparator output of 1's and 0's multiplied by the ADC referencevoltage will represent the signal level.

A cyclic ADC 1700 is shown in FIG. 17A. The basic operations aremultiplying the integrated value by 2 if the value is greater than theADC comparator reference level at input 1720 or multiply the integratedvalue by 2 and add the ADC reference if it is less than the ADCcomparator reference level at input 1720. A unity gain buffer 1790 isused to level shift the output of the summing node by 1V to allow thePPD_1 devices to fill up with electrons in proportion to the DAC output.The multiply by 2 operation is achieved by setting the number of PPD_1devices in the charge transfer stages 1740 and transfer operations usedto spill electrons into the summing node 1708. The subtraction of thereference level is achieved by multiple HPD_1 device transfers from thecharge subtractor stages 1750 used to spill holes into the summing node1708.

The cyclic ADC shown in FIG. 17A samples the input into the modulatorthrough the select switches 1722-1 and 1722-2 with timing shown on FIG.17B. In particular, the switch 1722-1 is used to sample the an inputprovided by pixel buffer 1301. When the switch 1722-1 is used to connectthe pixel buffer 1301 to the integrator 1740, the “sample_input” signalmay be asserted as shown in FIG. 17B. The switch 1722-2 may be used tosample the feedback from the unity gain level shifting buffer 1790, andis used to connect the output of buffer 1790 to the integrator 1740 whenthe “sample_feedback_from_integrator” signal is asserted in FIG. 17B.Conversion of the pixel signal level only is shown (0.6V signal belowthe reference). Operation of the cyclic ADC is based on firstdetermining the MSB to see if the signal level magnitude is greater thanthe ADC reference voltage divided by 2. The integrator is first reset to2.5V that represents the level with 0 signal in event 1780. Theintegrator output signal swing has a maximum of 1V down to 1.5V from2.5V. The comparator reference is set at 2.0V that represents the halfthe ADC reference level (1V). The cyclic algorithm operates such thatafter the MSB is determined the result is subtracted from the integrator(0 signal if the MSB is 0 and half the reference if the MSB is 1). Inorder to determine the next bit, the remaining residue is compared tothe MSB-1 comparator reference level (Vref/4). In order to re-use thesame reference level used for the MSB bit, the residue is insteadmultiplied by 2 to effectively determine the MSB-1 bit. As a result theoverall equivalent operation is to multiply the residue by 2 andsubtract the ADC reference level (2×Vref/2=Vref).

After the initial pixel input signal is sampled through activation ofFill1 to Fill32 such as event 1781, the comparator is strobed in event1782. Then, the integrator output is sampled into the PPD bank with 2×the number of PPDs used to double the integrator value. This integratorsample process uses the signal Fill1x to Fill64x (2 times as many Fillsignals as used for the initial sample of the input) as shown in events1783-1 for charge transfer stages 1 to 32 and 1783-2 for charge transferstages 33 to 64. The value is held in the PPD as the Cdac integratorcapacitor is preset back to 2.5V in event 1784, which occurs between theassertion of the fill signals in events 1783-1 and 1783-2 and theassertion of the transfer signals in events 1783-1 and 1783-2. Then, thePPD values are transferred using the TX1 to TX64 signals by assertingthe transfer signals in events 1783-1 and 1783-2. If the integratoroutput is below the comparator threshold from the previous strobeoperation, the hole based HPD PPD adds charge to the Cdac node to movethe Cdac node by the ADC reference voltage by pulsing the Fill1h/TX1h toFill64h/TX64h signals.

In the timing diagram, the Cdac node is shown with the ADC referencebeing added first and then the 2x signal level charge being subtractednext but these can happen simultaneously.

The timing diagram shows the example of a signal level that is 0.6Vbelow the reference level (1.5V). The first 5 bit conversions for theADC are shown but additional bits of resolution are possible.

Because the circuits that sample the input signal level into theintegrator need to have a linear relationship between the input voltageand output voltage over the entire range of input signal value, apre-emphasis circuit is needed to compensate for the PPD non-linearcharge handling capacity. The circuit of FIGS. 18 and 19 both compensatefor this non-linearity. The circuit FIG. 18 compensates for signals thatonly add electrons to the Cdac node for an always decreasing signallevel. The circuit of FIG. 19 compensates for signals that can addpositive signal or negative signal to the Cdac node relative to a commonmode for either an increasing or decreasing signal level. The firstexample of the pre-emphasis circuit uses a sensing node Vsense tofeedback to the input of a switched capacitor amplifier to modulate theamount of charge transferred to the sensing node to ensure the outputachieves the target voltage change. The amplifier compares the sensenode output to the input voltage to adjust the amplifier output that isdrives the input to the PPD during the fill phase. Then, this charge istransferred to the capacitor on the C sense node. This step is repeatedmultiple times (e.g., 4 to 5 times). As charge is transferred to thesense node, the same operation is happening in parallel on theaccumulator summing node that resides in the target circuit (ADC) thatneeds the pre-emphasis input correction. When the sense node reaches thetarget voltage change, the output of the switch capacitor amplifier goesto a high enough voltage to not create charge in the PPD.

At the start of the pre-emphasis operation, the C sense node isprecharged to 2.5V and the amplifier circuit is auto-zeroed to set theproper operating voltage levels. Then, the switches to the amplifier forautozero are turned off and the Pixel output input signal is connectedto the amplifier positive input and the output of the amplifier isconnected to the feedback capacitor (2.5 fF in this example). The “zero”signal level for the input to the amplifier is 1.5V. The amplifieroutput is also set to 1.5V from the auto-zero operation. When theFill_sense signal is turned on, this 1.5V level will not inject chargeinto the PPD because it equals the PPD Vpin voltage. However, for signallevels that drop below 1.5V that are input to the amplifier through thesample switch, the output of the amplifier will drop to a low voltage.In the timing diagram, the input signal swing is 0.3V for the Pixeloutput at 1.2V relative to the 1.5V zero signal reference.

The amplifier output initially drops by twice the input signal or 0.6Vfor the topology shown and this gain in signal magnitude is set by the“feedback” factor in the circuit. The “feedback” factor is determined bythe size of the feedback capacitor and the total capacitance on thenegative input to the amplifier. Because the feedback capacitor is 2.5fF and the overall capacitance of the negative input to the amplifier isin the schematic is approximately 2.5 fF (assuming other capacitanceassociated with the circuit connected to the negative input node aremuch smaller), only half of the amplifier output signal is fed back tothe negative input. Because the feedback will force the negative inputto equal the positive input to the amplifier, the amplifier output gainsthe signal by 2 with the Voutp changing from 1.5V to 0.9V (0.6V change).Thus, the circuit amplifies the input signal by 2 as charge isaccumulated on the Csense node but there is no amplification of thesignal in the accumulator (gain=1) because the Cdac1 node is sized tocompensate for this gain in the sense circuit (Cdac1 is size 2× biggerthan Csense).

When the amplifier output is driven into the PPD _1 by turning on theFill_sense and the resulting charge is transferred to the Csensecapacitor from the TX_sense gate turning on, it is shown in the timingdiagram that the Vsense node changes from 2.5V to 2.1V or by only 0.4V.Because the capacitor feedback from the Csense node to the amplifiernegative input is 2.5 fF, the signal change initially is negative 0.2V(half of the 0.4V change). The amplifier output responds by changing bya positive 0.4V to equalize the negative amplifier input to the positiveamplifier input. As shown in the timing diagram, the Voutp signal movesup from 0.9V by 0.4V to 1.3V. Then, this output level is sampled againby the PPD and charge is transferred to the Csense node. Because thesignal being sampled into the PPD node is closer to the Vpin voltage,less charge is ultimately transferred to the Csense node. In this step,the signal change is only 0.1V on the Csense node as it moves down from2.1V to 2.0V. During the feedback process again, the amplifier outputchanges by 0.1V from 1.3V to 1.4V. These steps continue for a fixednumber of cycles and the Csense node approaches the target value of 0.6Vbelow the initial level of 2.5V. The amplifier output is also drivingthe accumulator input in the target circuit to achieve a linearlyproportional amount of charge transfer on the Cdac1 node. Thus thesignal input into the accumulator is linearly proportional to the inputvoltage to the circuit.

The circuit of FIG. 19 allows positive or negative signals to be addedto an accumulator. The circuit is similar to the previous pre-emphasiscircuit but contains a sense node that is connected to both electronbased PPD and hole based HPD devices. Also, the accumulator that isdriven has both electron based PPD and hole based HPD devices.

The sense circuit is set for a common mode input voltage of 2.0V.Negative signals relative to this common mode level are allowed from2.0V to 1.5V. Positive signal relative to the common mode level areallowed from 2.0V to 2.5V. The top amplifier operates on the negativesignals between 2.0V and 1.5V. The bottom amplifier operates on thepositive signals between 2.0V and 2.5V. Most input signals will beprocessed by only the top or bottom amplifier but for signals close tothe input common mode it is expected that both may operate in parallel.Both top and bottom operating in parallel is not a problem becausefeedback will eventually allow the output on Csense to converge to thetarget voltage.

The common mode for the top amplifier that processes negative signalshas the amplifier auto-zero output voltage equal to 1.5V. Any negativeinput signals relative to the common mode voltage will push this voltageto a lower level and the PPD will fill with electrons. These electronsget transferred to the Csense node and feedback operates as in theprevious “single” polarity design.

The common mode for the bottom amplifier that process positive signalshas the amplifier auto-zero output voltage equal to 2.5V. Any positiveinput signal relative to the common mode voltage will push this voltageto a higher level and the hole based HPD will fill with positive holes.These holes will get transferred to the Csense node and feedbackoperates in a similar fashion as the electron based pre-emphasiscircuit.

Note that the Csense node is initially precharged to 2.0V and allowed toswing between 2.5V and 1.5V. This signal swing range still allowselectron or holes to be fully transferred from the PPD/HPD devices. Theintegrator in the target accumulator also has this same allowed signalswing.

The PPD based ADC circuits including single slope, SAR, delta sigma, andcyclic architectures require arrays of PPD devices for theiraccumulators. These PPD devices (or HPD devices) can be dedicateddevices that are configured for each ADC. Also, it is possible to timemultiplex the structures that are natively part of the imaging array.The diagrams shown on FIG. 20 include the pixel schematic with the pixelout, RST, RS, TX, AB, as well as power connections for the sourcefollower SF, reset transistor drain, and AB transistor drain. The powerconnections for the reset transistor drain and AB drain can bemultiplexed to allow different pixel modes to allow ADC circuitconfigurations. The reset transistor drain can be part of the integratorDAC node and is labeled Vdac_node. The AB transistor drain is the drainfor the fill operation of a PPD and is labeled Vfill_mode.

For a group of pixels these special connections to the pixel array canbe switched between being connected to 2.8V or to the circuitconnections for the Vdac_node as well as Vfill transistor node. Thedrawing on p. 10 shows a group of pixels called a “Panel” with an arrayof pixels in it. These global connections for Vdac_node and Vfill_nodeare shown in the drawing. Local to each group of pixels in the panel isa switch to connected the panel signals for either imaging mode or ADCmode. The Vdac_node and Vfill_node signals for a panel are connected toother parts of the ADC circuit as shown in previous schematics.Depending on the chip layout the remaining part of the circuit islocated in another part of the sensor or for stacking technology couldbe located on a die below the pixel array. The panels also can beconfigured to be a column of pixels, row of pixels, or a combination ofa certain number of rows and columns of pixels.

The schematic for the panel doesn't show the other pixel controls likeAB and TX (as well as row select, pixel output, or source followerpower) but these are present as well. The AB and TX controls can becontrolled on a per row basis or locally controlled for each panel(controls from the ADC control circuit or from the sensor imaging modecontroller). Also, the RST gate for the reset transistor would be biasedon to allow the FD node that is part of the pixel to be connected to theADC Cdac node.

Note that each ADC can utilize both a combination of pixel placed PPDcircuits and also a bank of dedicated PPD circuits. Then a balancebetween using dedicated area for the ADC PPD circuits and borrowing timefrom the imaging mode in the pixel array can be made to optimize thesensor area, speed, and maximum imaging integration time. Underconditions when the pixel array integration time is short, there issignificant time when the pixel is not being used for imaging (dead timebetween exposures). Under these conditions, the ADC does not need toborrow integration time from the sensor operation. Also, it should benoted that there could be power benefits in operating the ADC withdevices from the pixel array because some of the charge used to fill upthe PPD device is generated by locally in the pixel from incoming light.

Note that any of the circuit configurations can be achieved with 1 to“n” number of PPD devices to tradeoff circuit parameters like size vs.speed. The capacitor size for the Cdacs can also be optimized along withthe PPD charge capacity. Using a single PPD but doing multiple “fill”and “transfer” cycles achieves the same functional result as using aparallel set of PPD to perform more charge transfer in a given timeinterval.

Notably, both electron and hole based PPD charge packet cells can beused to decrease voltages or increase voltages on circuit nodes. Thiscapability can be used as a low power method to set voltages in circuitsbecause of the relatively small amount of charge moved around in thecircuit. Hence, this circuit technique could be used in any circuit withhigh impedance nodes.

This PPD charge packet circuit has the ability to harvest light energyto create the charge needed in the charge packet PPD cells rather thanusing “fill” transistors. The small amount of charge used by the cellsand the ability to use electron hole pairs generated by light make thesecircuits ideal for ultra-low power applications. Using this method manyof the pixel array circuits can be operated in ultra-low power modes anduse light energy to supply charge. Also, the charge used in the PPD canbe thermally generated by dark current that is proportional to the chipjunction and ambient temperature.

The foregoing is merely illustrative of the principles of this inventionand various modifications can be made by those skilled in the artwithout departing from the scope and spirit of the invention.

What is claimed is:
 1. A method of operating an image sensor thatincludes a pixel with an output line, the method comprising: with firstcharge transfer circuitry that includes at least a first pinnedphotodiode, transferring a first amount of charge to a first capacitivenode that is coupled to a first input of a comparator; with a samplingtransistor, sampling a pixel signal onto a second capacitive node thatis coupled to a second input of the comparator; and with second chargetransfer circuitry that includes at least a second pinned photodiode,transferring a second amount of charge to the second capacitive node. 2.The method defined in claim 1, further comprising: comparing voltages atthe first and second inputs of the comparator; and transferring a thirdamount of charge to the first capacitive node with the first chargetransfer circuitry in response to determining that the voltage at thefirst input of the comparator is greater than the voltage at the secondinput of the comparator, wherein the second charge transfer circuitry isused to transfer the second amount of charge in response to determiningthat the voltage at the first input of the comparator is less than thevoltage at the second input of the comparator.
 3. The method defined inclaim 2, wherein transferring the third amount of charge comprises:transferring half of the first amount of charge to the first capacitivenode.
 4. The method defined in claim 1, wherein transferring the firstamount of charge comprises: performing a first number of chargetransfers with the first charge transfer circuit, wherein each of thefirst number of charge transfer transfers an amount of charge based on afull-well capacity of the first pinned photodiode in the first chargetransfer circuit.
 5. The method defined in claim 4, wherein transferringthe second amount of charge comprises: performing a second number ofcharge transfers with the first charge transfer circuit, wherein thesecond number is less than the first number.
 6. The method defined inclaim 4, further comprising: with at least a first plurality of chargetransfer circuits, performing the first number of charge transfers witheach of the charge transfer circuits in the first plurality of chargetransfer circuits while the first charge transfer circuit istransferring charge.
 7. The method defined in claim 1, whereintransferring the second amount of charge comprises: transferring half ofthe first amount of charge to the second capacitive node.
 8. The methoddefined in claim 1, wherein sampling the pixel signal comprises:receiving a first voltage level from the pixel output line at a levelshifting circuit; and with the level shifting circuit, shifting thefirst voltage level to a second voltage level that is greater than thefirst voltage; and providing the shifted voltage to the samplingtransistor.
 9. The method defined in claim 1, wherein the first chargetransfer circuitry is coupled to a first fill voltage supply, andwherein the second charge transfer circuitry is coupled to a second fillvoltage supply, the method further comprising: adjusting the first andsecond fill voltage supply values while using the first and secondcharge transfer circuitry for charge transfer operations.
 10. The methodof claim 1, wherein the first charge transfer circuitry comprises afirst plurality of charge transfer stages, the method furthercomprising: performing a plurality of charge transfer operations,wherein each of the plurality of charge transfer operations usedifferent subsets of the first plurality of charge transfer stages. 11.The method of claim 1, the method further comprising: performing aplurality of charge transfer operations, wherein each of the pluralityof charge transfer operations transfers a different amount of charge tothe first capacitive node.